Dual mode transmitter-receiver and decoder for RF transponder tags

ABSTRACT

An enhanced backscatter RF-ID tag reader system and multiprotocol RF tag reader system is provided. In a multiprotocol mode, the system emits a non-stationary interrogation signal, and decodes a phase modulated backscatter signal by detecting a stronger phase component from quadrature phase representations or determining phase transition edges in a phase of a received signal. The RF tag reader system predicts or follows the phase of the backscatter signal, thereby avoiding interference from nulls in the received signal waveform due to the non-stationary interrogation signal, relative movement or environmental effects. An acoustic RF-ID tag detection system detects the reradiated signal corresponding to respective transformation of a signal in the tag. Detection of either type of RF-ID tags therefore is possible, and the absence of any tag or absence of any valid tag also determined.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of application Ser. No. 10/901,743,filed on Jul. 28, 2004 now U.S. Pat. No. 7,741,956, which is acontinuation of application Ser. No. 10/463,364, filed on Jun. 17, 2003,now U.S. Pat. No. 6,950,009, which is a continuation of application Ser.No. 10/150,849, filed on May 17, 2002, now U.S. Pat. No. 6,580,358,which is a continuation of application Ser. No. 09/641,649, filed onAug. 18, 2000, now U.S. Pat. No. 6,433,671, which is a continuation ofapplication Ser. No. 08/914,285, filed on Aug. 18, 1997, now U.S. Pat.No. 6,107,910, which claims the benefit of provisional application No.60/033,212, filed on Nov. 29, 1996.

FIELD OF THE INVENTION

The present invention relates to an RF-ID receiver system which iscompatible with both surface acoustic wave and semiconductor memorybased RF-ID tags, thereby allowing multiple RF-ID tag type environmentsto exist.

BACKGROUND OF THE INVENTION

A number of different schemes are known for encoding, transmitting anddecoding identification signals from RF-ID tags. However, these schemesare generally incompatible, therefore requiring proprietary readers toaccept encoded transmissions from tags of the same vendor. Even wherethe transmission scheme is not proprietary, there is no standardizationin the various RF-ID applications.

These RF-ID tags comprise, at a minimum, an antenna and a signaltransforming device, some known devices are very complex. There are twoparticular types of passive RF-ID tags which are used. A first typeincludes an electronic circuit, e.g., CMOS, to store digital ID datawhich is then modulated onto a received signal by means of an RFcircuit, e.g., a GaAs MESFET, transistor or controlled diode. Power forthe data storage and modulating circuit may be derived from aninterrogating RF beam or another power source, and power for thetransmission itself is also derived from the beam. In this type ofsystem, the interrogating RF beam is generally of fixed frequency, withthe resulting modulated signal at the same or a different frequency,with AM, FM, PSK, QAM or another known modulation scheme employed. Inorder to provide separation between the received and transmittedsignals, the modulated output may be, for example, a harmonic of theinterrogating RF beam. Such a system is disclosed in U.S. Pat. No.4,739,328, expressly incorporated herein by reference.

A known RF-ID interrogation system provides an interrogation signalwhich incorporates phase diversity, i.e., a phase which periodicallyswitches between 0° and 90° so that a null condition is not maintainedfor a period which would prevent RF-ID tag readout with a homodynereceiver. See, U.S. Pat. No. 3,984,835, incorporated herein byreference.

Likewise, a known system, described in U.S. Pat. No. 4,888,591,incorporated herein by reference, provides a semiconductor memory tagwhich is interrogated with a direct sequence spread spectrum signal,which allows discrimination of received signals based on signal returndelay. By employing a direct sequence spread spectrum having adecreasing correlation of a return signal with the interrogation signalas delay increases, more distant signals may be selectively filtered.This system employs a homodyne detection technique with a dual balancedmixer.

A second type of RF-ID tag includes a surface acoustic wave device, inwhich an identification code is provided as a characteristic time-domainreflection pattern in a retransmitted signal, in a system whichgenerally requires that the signal emitted from an exciting antenna benon-stationary with respect to a signal received from the tag. Thisensures that the reflected signal pattern is distinguished from theemitted signal. In such a device, received RF energy, possibly withharmonic conversion, is emitted onto a piezoelectric substrate as anacoustic wave with a first interdigital electrode system, from whence ittravels through the substrate, interacting with reflector elements inthe path of the wave, and a portion of the acoustic wave is ultimatelyreceived by the interdigital electrode system and retransmitted. Thesedevices do not require a semiconductor memory. The propagation velocityof an acoustic wave in a surface acoustic wave device is slow ascompared to the free space propagation velocity of a radio wave. Thus,assuming that the time for transmission between the radio frequencyinterrogation system is short as compared to the acoustic delay, theinterrogation frequency should change such that a return signal having aminimum delay may be distinguished, and the interrogation frequencyshould not return to that frequency for a period longer than the maximumacoustic delay period. Generally, such systems are interrogated with apulse transmitter or chirp frequency system.

Systems for interrogating a passive transponder employing acoustic wavedevices, carrying amplitude and/or phase-encoded information aredisclosed in, for example. U.S. Pat. Nos. 4,059,831; 4,484,160;4,604,623; 4,605,929; 4,620,191; 4,623,890; 4,625,207; 4,625,208;4,703,327; 4,724,443; 4,725,841; 4,734,698; 4,737,789; 4,737,790;4,951,057; 5,095,240; and 5,182,570, expressly incorporated herein byreference. The tags interact with an interrogator/receiver apparatuswhich transmits a first signal to, and receives a second signal from theremote transponder, generally as a radio wave signal. The transponderthus modifies the interrogation signal and emits encoded informationwhich is received by the interrogator/receiver apparatus.

Because the encoded information normally includes an identification codewhich is unique or quasi-unique to each transponder, and because thetransponders of this type are relatively light weight and small and maybe easily attached to other objects to be identified, the transpondersare sometimes referred to as “labels” or “tags”. The entire system,including the interrogator/receiver apparatus and one or moretransponders, which may be active or passive, is therefore oftenreferred to as a “passive interrogator label system” or “PILS”.

Other passive interrogator label systems are disclosed in the U.S. Pat.Nos. 3,273,146; 3,706,094; 3,755,803; and 4,058,217.

In its simplest form, the systems disclosed in these patents include aradio frequency transmitter capable of transmitting RF pulses ofelectromagnetic energy. These pulses are received at the antenna of apassive transponder and applied to a piezoelectric “launch” transduceradapted to convert the electrical energy received from the antenna intoacoustic wave energy in the piezoelectric material. Upon receipt of apulse, an acoustic wave is generated within the piezoelectric materialand transmitted along a defined acoustic path. This acoustic wave may bemodified along its path, such as by reflection, attenuation, variabledelay, and interaction with other transducers.

When an acoustic wave pulse is reconverted into an electrical signal itis supplied to an antenna on the transponder and transmitted as RFelectromagnetic energy. This energy is received at a receiver anddecoder, preferably at the same location as the interrogatingtransmitter, and the information contained in this response to aninterrogation is decoded. The tag typically has but a single antenna,used for both receiving the interrogation pulse and emitting aninformation bearing signal.

In general, a passive interrogator label system includes an“interrogator” for transmitting a first radio frequency signal; at leastone transponder which receives this first signal, processes it and sendsback a second radio frequency signal containing encoded information; anda receiver, normally positioned proximate to or integrated with theinterrogator, for receiving the second signal and decoding thetransponder-encoded information.

Known technologies allow separate interrogation systems to operate inclose proximity, for example by employing directional antennas andemploying encoded transmissions, such as a direct sequence spreadspectrum signal, which has reduced self-correlation as relative delayincreases, thus differentiating more distant signals.

In known passive transponder systems, the encoded information isretrieved by a single interrogation cycle, representing the state of thetag, or obtained as an inherent temporal signature of an emitted signaldue to internal time delays.

In the acoustic wave tags described above, the interrogator transmits afirst signal having a first frequency that successively assumes aplurality of frequency values within a prescribed frequency range. Thisfirst frequency may, for example, be in the range of 905-925 MHz,referred to herein as the nominal 915 MHz band, a frequency band thatmay be available. The response of the tag to excitation at any givenfrequency is distinguishable from the response at other frequencies.Further, because the frequency changes over time, the received responseof the tag, delayed due to the internal structures, may be at adifferent frequency than the simultaneously emitted signal, thusreducing interference.

Passive transponder encoding schemes include control over interrogationsignal transfer function H(s), including the delay functions f(z). Thesefunctions therefore typically generate a return signal in the same bandas the interrogation signal. Since the return signal is mixed with theinterrogation signal, the difference between the two will generallydefine the information signal, along with possible interference andnoise. By controlling the rate of change of the interrogation signalfrequency with respect to a maximum round trip propagation delay,including internal delay, as well as possible Doppler shift, the maximumbandwidth of the demodulated signal may be controlled.

The following references are hereby expressly incorporated by referencefor their disclosure of RF modulation techniques, transponder systems,information encoding schemes, transponder antenna and transceiversystems, excitation/interrogation systems, and applications of suchsystems: U.S. Pat. Nos. 2,193,102; 2,602,160; 2,774,060; 2,943,189;2,986,631; 3,025,516; 3,090,042; 3,206,746; 3,270,338; 3,283,260;3,379,992; 3,412,334; 3,480,951; 3,480,952; 3,500,399; 3,518,415;3,566,315; 3,602,881; 3,631,484; 3,632,876; 3,699,479; 3,713,148;3,718,899; 3,728,632; 3,754,250; 3,798,641; 3,798,642; 3,801,911;3,839,717; 3,859,624; 3,878,528; 3,887,925; 3,914,762; 3,927,389;3,938,146; 3,944,928; 3,964,024; 3,980,960; 3,984,835; 4,001,834;4,019,181; 4,038,653; 4,042,906; 4,067,016; 4,068,211; 4,068,232;4,069,472; 4,075,632; 4,086,504; 4,114,151; 4,123,754; 4,135,191;4,169,264; 4,197,502; 4,207,518; 4,209,785; 4,218,680; 4,242,661;4,287,596; 4,298,878; 4,303,904; 4,313,118; 4,322,686; 4,328,495;4,333,078; 4,338,587; 4,345,253; 4,358,765; 4,360,810; 4,364,043;4,370,653; 4,370,653; 4,388,524; 4,390,880; 4,471,216; 4,472,717;4,473,851; 4,498,085; 4,546,241; 4,549,075; 4,550,444; 4,551,725;4,555,618; 4,573,056; 4,599,736; 4,604,622; 4,605,012; 4,617,677;4,627,075; 4,641,374; 4,647,849; 4,654,512; 4,658,263; 4,739,328;4,740,792; 4,759,063; 4,782,345; 4,786,907; 4,791,283; 4,795,898;4,798,322; 4,799,059; 4,816,839; 4,835,377; 4,849,615; 4,853,705;4,864,158; 4,870,419; 4,870,604; 4,877,501; 4,888,591; 4,912,471;4,926,480; 4,937,581; 4,951,049; 4,955,038; 4,999,636; 5,030,807;5,055,659; 5,086,389; 5,109,152; 5,131,039; 5,144,553; 5,163,098;5,193,114; 5,193,210; 5,310,999; 5,479,160; and 5,485,520. In addition,foreign patents CH346388; DE1295424; DE2926836; DE969289; EP0207020;FR22601 15; GB1 130050; GB1 168509; GB1 187130; GB2103408; GB2247096;GB774797; GB987868; JP0138447; JP0189467; JP1 16054; JP5927278; andNE1566716, as well as the following references: “IBM TechnicalDisclosure Bulletin”, (vol. 20, No. 7; 12/77), pp. 2525-2526; “IEEETransactions on Vehicular Technology”, (vol. VT-26, No. 1), 2/77; p. 35;A. R Koelle et al. “Short-Range Radio-Telemetry for ElectronicIdentification using Modulated RF Backscatter”, by A. (Proc. Of IEEE,8/75; pp. 1260-1261).; Baldwin et al., “Electronic Animal . . .Monitoring”, 1973; Electronic Letters, December 1975, vol. 11, pp.642-643; Encyclopedia of Science and Technology; vol. 8, pp. 644-647(1982); Federal Information Processing Standards Publication 4A, Jan.15, 1977, Specifications for the Data Encryption Standard; IEEETransactions, Henoch et al., vol. MTTT-19, No. 1, January 1971; IEEETransactions, Jaffe et al., pp. 371-378, May 1965; IRE Transactions,Harrington, pp. 165-174, May 1962; IRE Transactions, Rutz, pp. 158-161,March 1961; J Lenk, Handbook of Microprocessors, Microcomputers andMinicomputers; p. 51 (1979); Koelle et al., “Electronic Identification .. . Monitoring”, 7/73 to 6/74, pp. 1-5; P. Lorrain et al., EMFields andWaves; Appendix A, (1970); Proceedings of IRE, March 1961, pp. 634-635;R. Graf, Dictionary of Electronics; p. 386, (1974); RCA Review, vol. 34,12/73, Klensch et al., pp. 566-579; RCA Review, Sterzer, 6/74, vol. 35,pp. 167-175; Reports on Research, September-October 1977, vol. 5, No. 2,each of which is expressly incorporated herein by reference.

SUMMARY AND OBJECTS OF THE INVENTION

The present invention provides a system providing a non-stationary radiofrequency emission and a receiver system capable of resolving both delaymodulation tags, e.g., surface acoustic wave tags, and state machinetags, e.g., semiconductor-based memory tags. The receiver must thereforedetermine a type of tag, if any, within an interrogation window, andsubsequently track a reradiated signal which is received simultaneously,and which is modulated both based on the emitted non-stationaryfrequency sequence and the internal modulation scheme, as well as areradiated signal which may be delayed in time.

Since the use of non-stationary radio frequency interrogation signalsand subsequent analysis of time domain delay modulated return signalcomponents is conventional, these known methods will not be exploredherein in detail. It is understood, however, that the present techniquemay be used to combine various different RF-ID techniques either in asingle hybrid tag system or in an environment with differing types oftags.

In addition, the present invention allows the use of spread spectrumtechnology to receive data from backscatter tags. Further, certaininteractive tags which download information from the interrogationsignal may also be compatible with the technique. In fact, since thenon-stationary sequence of the interrogation signal is normally ignoredby the tag, the sequence itself may be modulated to provide aninformation signal. The use of a non-stationary frequency forbackscatter tags is not heretofore known.

Typically, a state machine passive backscatter RF-ID tag provides anantenna which interacts with a received radio frequency signal. Amodulator alters the reflection or impedance characteristic of theantenna system, such that a backscatter signal which varies over time isemitted. The return signal is thus monitored for an informationtransmission protocol and a message extracted. Since these systemstypically are open loop, i.e., no feedback that a message has, in fact,been received by the receiver, redundant or continuous transmissions aremade. In order to increase the signal to noise ratio, the return signalis typically modulated using other than simple AM modulation. Where theexcitation signal is non-stationary, or the tag distant or moving withrespect to the transmitter or receiver, phase locking the receiver tothe transmitter may be ineffective as a demodulation scheme. Therefore,the present invention provides a system which tracks the modulationsignal of the backscatter signal while effectively ignoring signalcomponents, such as non-stationary frequency, movement induced Dopplereffects, and the like, which occur outside the symbol transition raterange of the tags.

Advantageously, in one embodiment, the receiver for a PSK modulated tagneed not operate in phase synchronous manner with the radio frequencycarrier. At the receiver, the signal from the tag is mixed with a signalwhich corresponds to the interrogation signal. A dual balanced mixergenerates outputs corresponding to both I and Q, however, the strongestphase is analyzed, and the weaker phase is ignored or analyzed, to theextent that it is expected to contain useful information. Because ofmany variables, the stronger phase may change many times during receiptof a message. In the case of more complex modulation schemes, it may benecessary to analyze the return signal more rigorously. However, thismay be accomplished using known signal analysis techniques.

Various advantages of spread spectrum communications may be obtainedaccording to the present invention. First, a receive may be sensitive tothe presence of interfering signals in the environment, especiallyfrequency stationary sources, and avoid employing these frequencies, orensuring that each tag is interrogated for each portion of a completecycle outside an interference scope. Further, by providing a common bandbroader than necessary for any one transponder system, a number ofinterrogation systems may share the same band and environment with lowrisk of interference. So long as the interrogation sequences arenon-overlapping, or uncorrelated, operation will be generally reliable,without need for coupling the interrogation systems. Of course, theinterrogation systems may also be coupled, to ensure that there islittle or no interference.

In a preferred embodiment of the invention, an excitation transmittedwaveform for detecting the reradiated radio frequency signal is a chirp,i.e., a signal which repetitively monotonically changes in frequencyover a range. For example, a sawtooth signal may provide an input to avoltage controlled oscillator. In this case, the phase of the chirpsignal is continuous, with a change in relative phase angle over timeuntil a limit is reached. In these systems, it is expected that therange of change in frequency is significant and the rate of frequencychange is high. Therefore, approximations which rely on a slowly varyingsignal or small range of variation are inapplicable.

The chirp signal, derived from the excitation signal source, is mixedwith a local oscillator signal for downconversion, generating I and Qintermediate frequency (IF) signals. The IF signals are, in turn,detected with an AM detector. In this case, the phase of the IF signalis not stable with respect to the local oscillator, and thus the signalpower will migrate between the I and Q phases. Therefore, the preferredembodiment analyzes both the detected I and Q signal, to determine thedata encoded on the received waveform. For example, the stronger signalmay be presumed to have the signal with the highest signal to noiseratio, and therefore used exclusively in the signal analysis. The I andQ signals may also be analyzed together. Since the phase is presumed tobe instable, and in fact may be rotating, the stronger of the I and Qsignals will oscillate.

This method may also have applicability to other types of modulationschemes which do not employ quadrature phase modulation techniques,e.g., QAM, such that any one phase of the demodulated signal includesall of the modulated information of interest.

By allowing a modulated backscatter radio-frequency identification tagsto coexist in an environment with surface acoustic wave identificationtags, the present invention simplifies system operation with differingtag types and allows a system to be established with a future change inpreferred tag type, without redundant or incompatible equipment.

In order to read a known type of SAW RF-ID tag, e.g., an acoustictransponder available from XCI, Inc, San Jose, Calif., a non-stationaryfrequency radio frequency interrogation signal is transmitted to thetag, where it is modified by the SAW device, such as by reflectingand/or delaying portions of the wave so that a return signal ismodulated. In environments including multiple tag types, the type of tagis typically unknown until a response is received. Therefore, it is anobject according to the present invention to accept and decode a returnsignal from a semiconductor memory RF-ID tag from irradiation with afrequency modulated radio frequency interrogation beam, while alsoaccepting and decoding a response from a surface acoustic wave RF-IDtag, and determining a tag type and encoding, or an absence of a validtag. For example, the absence of tag detector includes an eventdetector, e.g., a car in a toll lane, in conjunction with no output fromdecoders for the different types of transponder supported. Optionally,the system may determine the validity of a code, so that an invalid codemay be distinguished from an absent code, with possible differentprocessing.

Of particular note in the present invention, the “carrier” frequency isnot stationary, and therefore the receiver is capable of receivingdigitally modulated backscatter signals which are immediately modulatedand retransmitted, without substantial delay, as well as reradiatedradio frequency signals in which the encoding is presented as one ormore substantial delays of a retransmitted derivative of the excitationsignal, e.g., from a SAW-based RF tag. In the latter case, by retuningthe excitation signal periodically or continuously, the delayed signalsmay be detected, which would be difficult or impossible if theexcitation signal remained at the same frequency. On the other hand,this non-stationary excitation signal requires compensation before adigitally modulated backscatter signal may be detected. Thus, thepresent invention provides a multiprotocol tag reader, allowingdifferent types of tags to be reliably identified.

In acoustic RF-ID transponder systems, the information code associatedwith and which identifies the passive transponder is built into thetransponder at the time that a layer of metallization is fully definedon the substrate of piezoelectric material. This metallization thusadvantageously defines the antenna coupling, launch transducers,acoustic pathways and information code elements, e.g., reflectors. Thus,the information code in this case is non-volatile and permanent. Theinformation is present in the return signal as a set of characteristicperturbations of the interrogation signal, such as delay pattern andattenuation. In the case of a tag having launch transducers and avariable pattern of reflective elements, the number of possible codes isN×2^(M) where N is the number of acoustic waves launched by thetransducers and M is the number of reflective element positions for eachtransducer. Thus, with four launch transducers each emitting twoacoustic waves, and a potential set of eight variable reflectiveelements in each acoustic path, the number of differently codedtransducers is 2048. Therefore, for a large number of potential codes,it is necessary to provide a large number of launch transducers and/or alarge number of reflective elements. However, power efficiency is lostwith increasing device complexity, and a large number of distinctacoustic waves reduces the signal strength of the signal encoding theinformation in each. The transponder tag thus includes a multiplicity of“signal conditioning elements”, i.e., delay elements, reflectors, and/oramplitude modulators, are coupled to receive the first signal from atransponder antenna. Each signal conditioning element provides anintermediate signal having a known delay and a known amplitudemodification to the first signal. Where the signal is split intomultiple portions, it is advantageous to reradiate the signal through asingle antenna Therefore, a single “signal combining element.” coupledto all of the signal conditioning elements and/or signal portions isprovided for combining the intermediate signals to produce the secondsignal. This second signal is coupled out to the same or a separateantenna for transmission as a reply. As described above, the signaldelay elements and/or the signal combining element impart a known,unique informational code to the second signal.

Preferably, the passive acoustic wave transponder tag includes at leastone known (control) element, which assists in synchronizing the receiverand allows for temperature compensation of the system. As thetemperature rises, the piezoelectric substrate may expand and contract,altering the characteristic delays and other parameters of the tag.Although propagation distances are small, an increase in temperature ofonly 20° C. can produce an increase in propagation time by the period ofone entire cycle at a transponder frequency of about 915 MHz. Theacoustic wave is often a surface acoustic wave, although bulk acousticwave devices may also be constructed.

The receiving and decoding apparatus associated with the system includesapparatus for receiving the second signal from the transponder and amixer arranged to receive both the first signal and the second signalfor performing four quadrant multiplication of these two signals. Themixer is preferably a complex mixer, generating I and Q phases 90°apart, although the mixer may be polyphasic (having two, three or morephases) which may be symmetric or asymmetric. The difference frequencies(or frequencies derived from the difference frequencies) of the firstand second signals, respectively are then processed by one or moresignal processors, to detect the phases and amplitudes of the respectivedifference frequencies contained in the third signal in the case of anacoustic transponder, or a sequence of modulation states in the case ofa semiconductor modulator backscatter transponder, to determine theinformational code associated with the interrogated transponder. Wherethe code is provided as a set of time delays, the signal processorperforms a time-to-frequency transform (Fourier transform) on thereceived signal, to assist in determination of the various delayparameters. The characteristic delays (and phase shifts) of thetransducer then appear in the transformed data set at the receiver assignal energy having a time delay. Alternately, a set of matched filtersmay be implemented, and the outputs analyzed. Where the code is providedas a sequence of symbols, a time domain analysis will generally suffice.The preferred embodiment of the invention employs separate analyzercircuitry for differing encoding schemes, but the circuitry and analysismay also be consolidated into a single system, for example a digitalsignal processing scheme.

In practice, a passive interrogator label system is frequentlyconfigured such that a plurality of transponders are interrogated from anumber of locations. For example, if the transponders (labels) arecarried on persons who are authorized entry into a building, thetransmitting and receiving antennas are normally located near severaldoors to the building. According to the present invention, the signalanalysis of both acoustic and semiconductor based transponders may beremote from the interrogation antenna system.

As another example, the labels may be placed on cattle which aremonitored at a number of locations, such as a holding area, a feedingarea and the like. The labels may also be placed on railroad cars topermit car identification at various locations throughout a switchyardor rail network. Other uses of such systems are known, and in fact thewidespread acceptance of interrogation systems, be they passive oractive, have generated the problem addressed by the present invention,namely, the presence of many competing and incompatible standards.

Thus, the processing of the transponder signal may be divided betweenthe interrogator-transponder communication in the radio frequency range,and the decoding of the received information, with the two functionspotentially separated. The decoding system electronics may bemultiplexed to effectively service a number of locations efficientlythrough a network.

OBJECTS OF THE INVENTION

It is also an object of the invention to provide a versatile receiverwhich can extract additional information from a return signal andselectively communicate with a plurality of RF-ID tags simultaneously,e.g., a first semiconductor memory tag and a second SAW reflectorpattern memory tag.

It is a further object according to the present invention to provide atag which includes both semiconductor memory and electrode patternmemory.

It is also an object of the invention to provide a method forinterrogating a backscatter generating tag, comprising the steps of (a)generating an interrogation signal having a frequency within aninterrogation band; (b) emitting the interrogation signal as a radiowave signal; (c) interacting the emitted radio wave signal with abackscatter generating tag; (d) receiving a radio frequency backscattersignal from the tag; (e) mixing the received backscatter signal with aplurality of representations of the interrogation signal, each of saidplurality of representations differing in phase, to produce a pluralityof mixed signals; (f) comparing a respective signal strength of saidplurality of mixed signals; and (h) analyzing said difference signalsover time to determine a significant information sequence of thebackscatter signal, while discounting an importance of at least one ofthe plurality of mixed signals at any given time based on said comparedrespective signal strengths.

It is a further object of the invention to provide a dual mode tagidentification system, in which a reradiated representation of aninterrogation signal is analyzed in a first mode to determine a transferfunction for said interrogation signal and in a second mode to determinea time sequence of modulation states imposed on said interrogationsignal.

It is still another object of the invention to provide a transponderinterrogation system for interrogating a transponder which receives aradio frequency wave and emits a modified radio frequency wave,comprising an interrogation radio frequency wave generator, generating aradio frequency excitation pulse adapted for probing a plurality ofcharacteristic time-constants of a transponder and for communicatingwith the transponder, an antenna, for receiving the modified radiofrequency wave; a first decoder, for determining the plurality ofcharacteristic time-constants from the modified radio frequency wave;and a second decoder for determining a sequence of modulation statesfrom the modified radio frequency wave.

It is also an object of the present invention to provide a backscattertransponder interrogation system, comprising: (a) an input for receivinga backscatter signal from a backscatter transponder due to aninterrogation signal; (b) a multiphase mixer for mixing said receivedbackscatter signal and a representation of said interrogation signal toproduce multiphasic outputs; (c) means for selecting a mixer multiphasicoutput having substantial signal strength; and (d) a decoder fordecoding a sequence of symbols from the selected mixer multiphasicoutput.

It is another object of the present invention to provide an RF-ID taginterrogator, responsive to a return signal from an RF-ID tag having asemiconductor device outputting symbols which are accessed serially overtime to sequentially modulate an interrogation signal at a modulationrate, comprising: a transmitter, transmitting a radio frequencyinterrogation signal, said interrogation signal having a frequency whichsubstantially varies over time; a receiver, receiving a signal from theRF-ID tag which corresponds to said radio frequency interrogationsignal, sequentially modulated over time based on the symbols; adecoder, having: a phase-sensitive demodulator, for extracting a complexmodulation pattern from said received signal, with respect to arepresentation of said interrogation signal; a symbol detector receivingsaid complex modulation pattern, extracting a data clock from one ofsaid complex modulation pattern, said interrogation signal, or areference clock, compensating for a phase rotation in the complexmodulation pattern due to frequency variation of said interrogationsignal at a rate faster than the modulation rate, and extracting saidsymbols from said compensated complex modulation pattern and said dataclock.

The present invention also provides as an object a radio frequencyreceiving device, operating in an environment including an RF generator,generating a time-variant RF signal which propagates through space, andan RF signal modulator having a frequency modulation pattern based ondata symbols stored in said device, comprising: an input, receiving afrequency modulated signal corresponding to said time-variant RF signalmodulated by the data symbols; a demodulator, producing a demodulatedsignal by mixing a signal corresponding to said time-variant RF signalwith said received signal, while preserving a phase pattern; acomparator, selecting a phase component having a greatest magnitude fromat least two phase components having differing phase axes of saiddemodulated signal, said comparator having a magnitude selectivitypattern excluding selection of a component based primarily on a patternof said data symbols; a detector, detecting said selected phasecomponent to extract said data symbols; and an output, for outputtinginformation relating to said data symbols.

It is a further object of the invention to provide a device forreceiving information from a remote tag, the tag having informationstored in a memory and a modulator for frequency modulating an incidentsignal based on the stored information, comprising: a transmitter fortransmitting a radio frequency carrier having a time varying centerfrequency in proximity to the tag; a receiver for receiving a frequencymodulated, time varying center frequency carrier signal from the tag; abalanced mixer, receiving said frequency modulated, time varying centerfrequency carrier signal and said radio frequency carrier to produce atleast a difference signal with at least two outputs each representing adifferent phase axis; a detector circuit receiving said at least twooutputs and extracting the information from at least one of saidoutputs.

It is a still further object of the invention to provide a radiofrequency receiving device, operating in an environment including an RFgenerator, generating a phase-continuous, time-variant RF signal whichpropagates through space, and an RF signal modulator having a frequencymodulation based on data symbols, comprising: an input, receiving afrequency modulated signal corresponding to said time-variant RF signalmodulated by the data symbols; a demodulator, producing a demodulatedsignal by mixing a signal corresponding to said time-variant RF signalwith said received signal, while preserving a phase pattern; acomparator, selecting a phase component having a greatest magnitude fromat least two phase components having differing phase axes of saiddemodulated signal, said comparator having a magnitude selectivitypattern excluding selection of a component based primarily on a patternof said data symbols; a detector, detecting said selected phasecomponent to extract said data symbols; and an output, for outputtinginformation relating to said data symbols.

It is also an object of the present invention to provide an RF-ID tagreader, responsive a return signal from an RF-ID tag having an RF outputmodulating an interrogation signal over time in a pattern correspondingto a sequence of symbols, comprising: a receiver, receiving a modulatedsignal from the RF-ID tag which corresponds to said radio frequencyinterrogation signal, modulated over time based on the symbols; acomplex demodulator, for demodulating in complex space a modulatedsignal pattern of the received modulated signal to produce at least twophases and preferentially producing an output based on a phase having agreater signal strength, to extract a modulation pattern from saiddemodulated signal; an analyzer for reconstructing the symbols from thedetected modulation pattern; an output for producing informationcorresponding to said sequence of symbols.

It is stiff further object of the invention to provide a RF-ID tagbackscatter demodulator having a signal relative phase change detectorfor determining a relative phase change in a received signal.Preferably, quadrature phase representations of the signal are comparedwith respective delayed quadrature representations to detect a relativephase reversal edge, with analysis of the quadrature phase edge signalsbased on a quadrature phase received signal strength.

These and other objects will become apparent from a review of thedetailed description of the preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS AND APPENDICES

FIG. 1 is a block diagram of a known passive interrogator label system;

FIG. 2 is a block diagram of a transponder or “label” which is used inthe system of FIG. 1;

FIGS. 3A and 3B are time diagrams, drawn to different scales, of theradio frequencies contained in the interrogation and reply signalstransmitted with the system of FIG. 1.

FIG. 4 is a block diagram illustrating the decoding process carried outby the signal processor in the system of FIG. 1.

FIG. 5 is a block signal diagram of a passive transponder which may beused with the system of FIG. 1.

FIG. 6 is a plan view, in enlarged scale, of a first configuration ofthe transponder of FIG. 5.

FIG. 7 is a plan view, in greatly enlarged scale, of a portion of thetransponder configuration shown in FIG. 6.

FIGS. 8-12 are representational diagrams, in plan view, of a variousconfigurations of the transponder of FIG. 5.

FIG. 13 is a plan view, in enlarged scale, of a seventh configuration ofthe transponder of FIG. 5.

FIG. 14 is a plan view, in greatly enlarged scale, of a portion of thetransponder configuration shown in FIG. 13.

FIG. 15 is a diagram showing the respective time delays of the differentSAW pathways in the transponder of FIG. 13.

FIG. 16 is a flow diagram showing the order of calculations carried outby the signal processor and microprocessor in the system of FIG. 1.

FIG. 17 is a diagram illustrating SAW reflections from conventionaltransducer fingers.

FIG. 18 is a diagram illustrating SAW reflections from split transducerfingers, according to the invention.

FIG. 19 is a diagram illustrating the energy converted by a SAWtransducer at its fundamental (resonant) frequency and at its thirdharmonic.

FIG. 20 is a representational diagram, in plan view, of a launch and/orreceiving transducer of the type employed in the transponder of FIGS.5-14.

FIG. 21 is a diagram showing the operational bandwidth of the transducerof FIG. 13.

FIG. 22 is a representational diagram of two transducers connected inseries.

FIG. 23 is a representational diagram of a single transducer formed oftwo separate transducers connected in series.

FIG. 24 is a cross-sectional view, greatly enlarged, of a section of thetransducer illustrated in FIGS. 6 and 7.

FIG. 25 is an equivalent circuit diagram showing the resistance andcapacitance of a SAW transducer.

FIG. 26 is a plan view, in greatly enlarged scale, of a reflector andthree delay pads in the transducer configuration of FIGS. 6 and 7.

FIG. 27 is a plan view, in greatly enlarged scale, of the edge of adelay pad showing anti-reflection serrations.

FIG. 28 is a plan view, in greatly enlarged scale, of a reflector havingnumerous short circuits between the respective reflector fingers.

FIG. 29 is a schematic drawing of a processor circuit for analyzing apassive transponder signal.

FIGS. 30, 31 and 34 are simplified block diagrams of a reader and anactive transponder for identifying at the reader al object associatedwith the transponder.

FIG. 32 is a schematic drawing of a transmitter of FIG. 31.

FIG. 33 is a schematic drawing of a sequence generator of FIG. 32.

FIG. 35 shows various signals involved in the operation of a directsequence spread spectrum transponder interrogation system.

FIGS. 36-38 show distance relationships of the direct sequence spreadspectrum transponder interrogation system.

FIGS. 39 a, 39 b, 39 c and 39 d constitute complementary block diagramsillustrating on a somewhat detailed basis the construction of the readershown in FIG. 30.

FIG. 40 illustrates waveforms produced in the transponder and detectedat the reader to identify a binary “1”, a binary “0” and a codeindicating the end of the generation at the transponder of sequences ofsignal cycles in the pattern of binary 1's and binary 0's identifyingthe object.

FIG. 41 illustrates waveforms of signal cycles generated at the readerto detect the sequence of binary 1's and binary 0's identifying theobject.

FIG. 42 illustrates waveforms of signal cycles generated at the readerto produce phase-locked signals used at the reader to provide clocksignals for synchronizing the operation of the reader shown in FIGS. 39a, 39 b, 39 c, and 39 d.

FIG. 43 illustrates waveforms of additional signals generated at thereader to produce additional phase-locked signals for providing theclock signals.

FIGS. 44A-E show a schematic drawing of an active transponderinterrogation system, having an edge sensitive transition detector, andrespective signals during operation of the circuit.

FIG. 45 is a waveform and timing diagram for the various signalsreceived and generated by the EPLD according to FIG. 48;

FIGS. 46-49 show schematic drawings of portions of a multimode activetransponder interrogation system; and

Appendix A provides a source code listing according to the presentinvention for a decoding program for a backscatter transponder asdisclosed in U.S. Pat. No. 4,739,328.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The preferred embodiments of the present invention will now be describedwith reference to FIGS. 1-49 of the drawings. Identical elements in thevarious figures are designated with the same reference numerals.

Acoustic Wave Transponder Tag

A surface acoustic wave passive interrogator label system, as described,for example, in U.S. Pat. Nos. 4,734,698; 4,737,790; 4,703,327; and4,951,057, includes an interrogator comprising a voltage controlledoscillator 10 which produces a first signal S1 at a radio frequencydetermined by a control voltage V supplied by a control unit 12. Thissignal S1 is amplified by a power amplifier 14 and applied to an antenna16 for transmission to a transponder 20. The voltage controlledoscillator 10 may be replaced with other oscillator types.

The signal S1 is received at the antenna 18 of the transponder 20 andpassed to a signal transforming element 22. This signal transformerconverts the first (interrogation) signal S1 into a second (reply)signal S2, encoded with an information pattern. The information patternis encoded as a series of elements having characteristic delay periodsT₀ and ΔT₁, ΔT₂, . . . ΔT_(N). Two common types of systems exist. In afirst, the delay periods correspond to physical delays in thepropagation of the acoustic signal. After passing each successive delay,a portion of the signal I₀, I₁, I₂, . . . I_(N) is tapped off andsupplied to a summing element. The resulting signal S2, which is the sumof the intermediate signals I₀ . . . I_(N), is fed back to a transpondertag antenna, which may be the same or different than the antenna whichreceived the interrogation signal, for transmission to theinterrogator/receiver antenna. In a second system, the delay periodscorrespond to the positions of reflective elements, which reflectportions of the acoustic wave back to the launch transducer, where theyare converted back so an electrical signal and emitted by thetransponder tag antenna.

The signal S2 is passed either to the same antenna 18 or to a differentantenna 24 for transmission back to the interrogator/receiver apparatus.This second signal S2 carries encoded information which, at a minimum,identifies the particular transponder 20.

The signal S2 is picked up by a receiving antenna 26. Both this secondsignal S2 and the first signal S1 (or respective signals derived fromthese two signals) are applied to a mixer (four quadrant multiplier) 30to produce a third signal S3 containing frequencies which include boththe sums aid the differences of the frequencies contained in the signalsS1 and S2. The signal S3 is passed to a signal processor 32 whichdetermines the amplitude a_(i), and the respective phase Φ_(i) of eachfrequency component Φ_(i) among a set of frequency components Φ₀, Φ₁, Φ₂. . . ) in the signal S3. Each phase Φ_(i) is determined with respect tothe phase Φ₀=0 of the lowest frequency component Φ₀. The signal S3 maybe intermittently supplied to the mixer by means of a switch, and indeedthe signal processor may be time-division multiplexed to handle aplurality of S3 signals from different antennas.

The information determined by the signal processor 32 is passed to acomputer system comprising, among other elements, a random access memory(RAM) 34 and a microprocessor 36. This computer system analyzes thefrequency, amplitude and phase information and makes decisions basedupon this information. For example, the computer system may determinethe identification number of the interrogated transponder 20. This IDnumber and/or other decoded information is made available at an output38.

The transponder may be an entirely passive device, or it may contain apower source and one or more active elements.

The transponder serves as a signal transforming element 22, whichcomprises N+1 signal conditioning elements 40 and a signal combiningelement 42. The signal conditioning elements 40 are selectively providedto impart a different response code for different transponders, andwhich may involve separate intermediate signals I₀, I₁ . . . I_(N)within the transponder. Each signal conditioning element 40 comprises aknown delay T_(i) and a known amplitude modification A_(i) (eitherattenuation or amplification). The respective delay T_(i) and amplitudemodification A_(i) may be functions of the frequency of the receivedsignal S1, or they may provide a constant delay and constant amplitudemodification, respectively, independent of frequency. The time delay andamplitude modification may also have differing dependency on frequency.The order of the delay and amplitude modification elements may bereversed; that is, the amplitude modification elements A_(i) may precedethe delay elements T_(i) Amplitude modification A_(i) can also occurwithin the path T_(i).

The signals are combined in combining element 42 which combines theseintermediate signals (e.g., by addition, multiplication or the like) toform the reply signal S2 and the combined signal emitted by the antenna18.

In one embodiment, the voltage controlled oscillator 10 is controlled toproduce a sinusoidal RF signal with a frequency that is swept in 128equal discrete steps from 905 MHz to 925 MHz. Each frequency step ismaintained for a period of 125 microseconds so that the entire frequencysweep is carried out in 16 milliseconds. Thereafter, the frequency isdropped back to 905 MHz in a relaxation period of 0.67 milliseconds. Thestepwise frequency sweep 46 shown in FIG. 3B thus approximates thelinear sweep 44 shown in FIG. 3A.

Assuming that the stepwise frequency sweep 44 approximates an average,linear frequency sweep or “chirp” 47, FIG. 3B illustrates how thetransponder 20, with its known, discrete time delays T₀, T₁ . . . T_(N)produces the second (reply) signal 52 with distinct differences infrequency from the first (interrogation) signal 51. Assuming around-trip, radiation transmission time of t₀, the total round-triptimes between the moment of transmission of the first signal and themoments of reply of the second signal will be t₀+T₀, t₀+T₁, . . .t₀+T_(N), for the delays T_(0N), T . . . , T₁ respectively. Consideringonly the transponder delay T_(N), at the time t_(R) when the second(reply) signal is received at the antenna 26, the frequency 48 of thissecond signal will be Δf_(N) less than the instantaneous frequency 47 ofthe first signal S1 transmitted by the antenna 16. Thus, if the firstand second signals are mixed or “heterodyned”, this frequency differenceΔf_(N) will appear in the third signal as a beat frequency.Understandably, other beat frequencies will also result from the otherdelayed frequency spectra 49 resulting from the time delays T₀, T₁ . . .T_(N-1). Thus, in the case of a “chirp” waveform, the difference betweenthe emitted and received waveform will generally be constant.

In mathematical terms, we assume that the phase of a transmittedinterrogation signal is Φ=2πfτ, where τ is the round-trip transmissiontime delay. For a ramped frequency df/dt or f, we have: 2πfτ·=dΦ/dt=ω·ω,the beat frequency, is thus determined by τ for a given ramped frequencyor chirp f.

In this case, the signal S3 may be analyzed by determining a frequencycontent of the S3 signal, for example by applying it to sixteen bandpassfilters, each tuned to a different frequency, f₀, f₁ . . . f_(E), f_(F).The signal processor determines the amplitude and phase of the signalsthat pass through these respective filters. These amplitudes and phasescontain the code or “signature” of the particular signal transformer 22of the interrogated transponder 20. This signature may be analyzed anddecoded in known manner.

The transponder system typically operates in the band 905-925 MHz, with128 frequency steps. In a swept frequency embodiment, the sequence offrequencies are determined by noting successive constant frequencyincrements (Δf) above 905 MHz. Frequency changes are determined by adelay line detector comprising a delay element 86, a mixer 88, lowpassfilter 90 and a zero-crossing detector 92. The signal S1 is passedthrough the delay element 86 to one input of the mixer 88, and passeddirectly to the other input of the mixer 88. This mixer produces anoutput signal containing both the sum and difference frequencies of thetwo signals applied thereto. This output is supplied to the lowpassfilter 90 which passes only the portion of the signal containing thedifference frequencies. The output of the lowpass filter is supplied tothe zero-crossing detector 92 which produces a pulse at each positive(or negative) going zero-crossing. These pulses are passed to themicroprocessor 76 to inform the microprocessor when the frequency of thesignal S1 has changed by a fixed increment (Δf) of 156.25 KHz.

During normal operation of the interrogator apparatus, themicroprocessor 76 controls the frequency of the VCO 72 by successivelyretrieving the digital numbers from storage and supplying a differentnumber to the D/A converter 78 every 125 microseconds (i.e., at an 8 kHzrate). In actual operation the microprocessor inserts two additional,uniformly placed steps (in frequency and time) between the stepscalculated from the delay line. This is done to help eliminate “spectralaliasing” of the difference frequencies.

A clock 32 is provided having a 100 KHz output. As may be seen, thesignal 68 is received during the interval between transmissions of thesignal 66. These intervals are chosen to equal, approximately, the roundtrip delay time between the transmission of a signal to the transponderand the receipt of the transponder reply. The transponder reply willcontain a number of frequencies at any given instant of time as a resultof the combined (i.e., summed) intermediate signals having differentdelay times (T₀, T₀+ΔT, T₀+2ΔT, . . . T₀+NΔT).

In one embodiment of a passive transponder, shown in FIGS. 6 and 7, theinternal circuit operates to convert the received signal S1 into anacoustic wave and then to reconvert the acoustic energy back into anelectrical signal S2 for transmission via a dipole antenna 70, connectedto, and arranged adjacent a SAW device made of a substrate 72 ofpiezoelectric material such as lithium niobate. More particularly, thesignal transforming element of the transponder includes a substrate 72of piezoelectric material such as a lithium niobate (LiNbO₃) crystal,which has a free surface acoustic wave propagation velocity of about3488 meters/second. On the surface of this substrate is deposited alayer of metal, such as aluminum, forming a pattern which includestransducers and delay/reflective elements.

One transducer embodiment includes a pattern consisting of two bus bars74 and 76 connected to the dipole antenna 70, a “launch” transducer 78and a plurality of “tap” transducers 80. The bars 74 and 76 thus definea path of travel 82 for a surface acoustic wave which is generated bythe launch transducer and propagates substantially linearly, reachingthe tap transducers each in turn. The tap transducers convert thesurface acoustic wave back into electrical energy which is collected andtherefore summed by the bus bars 74 and 76. This electrical energy thenactivates the dipole antenna 70 and is converted into electromagneticradiation for transmission as the signal S2.

The tap transducers 80 are provided at equally spaced intervals alongthe surface acoustic wave path 82, as shown in FIG. 6, and aninformational code associated with the transponder is imparted byproviding a selected number of “delay pads” 84 between the taptransducers. These delay pads, which are shown in detail in FIG. 7, arepreferably made of the same material as, and deposited with, the busbars 74, 76 and the transducers 78, 80. Each delay pad has a widthsufficient to delay the propagation of the surface acoustic wave fromone tap transducer 80 to the next by one quarter cycle or 90° withrespect to an undelayed wave at the frequency of operation (in the 915MHz band). By providing locations for three delay pads betweensuccessive tap transducers, the phase f of the surface acoustic wavereceived by a tap transducer may be controlled to provide four phasepossibilities, zero pads=0°; one pad=90°; two pads=180°; and threepads=270°.

The phase information Φ₀ (the phase of the signal picked up by the firsttap transducer in line), and Φ₁, Φ₂ . . . Φ_(N) (the phases of thesignals picked up by the successive tap transducers) is supplied to thecombiner (summer) which, for example, comprises the bus bars 74 and 76.This phase information, which is transmitted as the signal S2 by theantenna 70, contains the informational code of the transponder.

As shown in FIG. 7, the three delay pads 84 between two tap transducers80 are each of such a width L as to each provide a phase delay of 90° inthe propagation of an acoustic wave from one tap transducer to the nextas compared to the phase in the absence of such a delay pad. This widthL is dependent upon the material of both the substrate and the delay paditself as well as upon the thickness of the delay pad and the wavelengthof the surface acoustic wave. As noted above, the substrate material ispreferably lithium niobate (LiNbO₃) and, the delay pad material ispreferably aluminum.

The transducers are typically fabricated by an initial metallization ofthe substrate with a generic encoding, i.e., a set of reflectors ordelay elements which may be further modified by removal of metal toyield the customized transponders. Thus, in the case of delay pads,three pads are provided between each set of transducers or taps, some ofwhich may be later removed. Where the code space is large, thesubstrates may be partially encoded, for example with higher order codeelements, so that only the lower order code elements need by modified ina second operation.

While a system of the type described above operates satisfactorily whenthe number of tap transducers does not exceed eight, the signal to noiseratio in the transponder reply signal is severely degraded as the numberof tap transducers increases. This is because the tap transducersadditionally act as launch transducers as well as partial reflectors ofthe surface acoustic wave so that an increase in the number of taptransducers results in a corresponding increase in spurious signals inthe transponder replies. This limitation on the number of taptransducers places a limitation on the length of the informational codeimparted in the transponder replies.

Spurious signals as well as insertion losses may be reduced in a passivetransponder so that the informational code may be increased in size toany desired length, by providing one or more surface acoustic wavereflectors on the piezoelectric substrate in the path of travel of thesurface acoustic wave, to reflect the acoustic waves back toward atransducer for reconversion into an electric signal.

A transducer 86 may thus be employed in conjunction with reflectors 88and 90 in a unique configuration which replaces the aforementionedarrangement having a launch transducer 78 and tap transducers 80. Inparticular, the transducer 86 is constructed to convert electricalenergy received at the terminals 92 and 94 into surface acoustic waveenergy which propagates outward in opposite directions indicated by thearrows 96 and 98. The launch transducer is constructed in a well knownmanner with an inter-digital electrode assembly formed of individualelectrode fingers arranged between and connected to the two bus bars 100and 102. In the illustrated pattern, half the fingers are connected tothe bus bar 100 and the other half are connected to the bus bar 102.Each electrode is connected to one or the other bus bar and extendstoward a free end in the direction of the other bus bar.

The distance between the centers of successive fingers is equal to 3λ/4where λ is the center wavelength of the surface acoustic wave.Furthermore, as may be seen, the length of the active region between theends of the electrodes connected to the bus bar 100 and the ends of theelectrodes connected to the bus bar 102 is Kλ, where K is aproportionality constant.

Surface acoustic waves which travel outward from the transducer 86 inthe directions 96 and 98 encounter and are reflected back by thereflectors 88 and 90. These reflectors comprise individual electrodefingers which extend between the bus bars 104 and 106 on opposite sides.These electrodes are spaced from center to center, a distance λ/2 apart.

The reflectors 88 and 90 serve to reflect nearly 100% of the surfaceacoustic wave energy back toward the transducer 86; that is, in thedirections 108 and 110, respectively. Thus, after a pulse of surfaceacoustic wave energy is generated by the transducer 86, it is reflectedback by the reflectors 88 and 90 and reconverted into an electricalsignal by the transducer 86.

The configuration may also include one or more delay pads 112 whichcontrol the phase of the surface acoustic wave received back by thetransducer 86. For a 90° phase delay (as compared to the phase of thereceived surface acoustic wave without a delay pad present) the delaypad should have a width equal to ½ the width of the typical delay padsbecause the surface acoustic wave will traverse the delay pads twice(i.e., in both directions).

A plurality of transducers 114 may be connected to common bus bars 116and 118 which, in turn, are connected to the dipole antenna of thetransponder. On opposite sides of this configuration and reflectors 120and 122 which reflect surface acoustic waves back toward the transducerswhich launched them.

Since the transducers 114 are connected in parallel, an interrogationpulse at radio frequency is received by, all the transducers essentiallysimultaneously. Consequently, these transducers simultaneously generatesurface acoustic waves which are transmitted outward in both directions.Due to the particular configuration shown, the reflected surfaceacoustic waves are received at staggered intervals so that a singleinterrogation pulse produces a series of reply pulses after respectiveperiods of delay.

Another embodiment of a passive transponder includes four transducers124 which are connected electrically in series between bus bars 126.These transducers are interconnected by means of intermediate electrodes128, the electrical circuit through each transducer being effected bycapacitive coupling. When energized by an RF electrical signal, thetransducers simultaneously produce surface acoustic waves which travelin four parallel paths 130.

To the right of the transducers 124 are four sets 132, 134, 136 and 138of reflectors 140 arranged in the paths of travel 130 of the surfaceacoustic waves. In the example shown, three reflectors 140 are arrangedin each set; however, the number of reflectors may be varied. If only asingle reflector is provided in each of the positions 132, 134, 136 and138, this reflector should be designed to reflect nearly 100% of thesurface acoustic waves at the wavelength of these waves. If more thanone reflector is provided, these reflectors should be designed toreflect only a portion of the acoustic wave energy.

Where three reflectors are provided in each set, the first and secondreflectors should allow some of the acoustic wave energy to pass beneaththem to the third and last reflector in line. In this way, if a pulse ofsurface acoustic wave energy is generated by a transducer 124, some ofit will be reflected by the first transducer, some by the second andsome by the third reflector in line.

Another transponder system provides separate launch and receivingtransducers. As may be seen, surface acoustic waves are generated by alaunch transducer 166 and propagated in the direction indicated by thearrow 168. These surface acoustic waves pass beneath the receivingtransducer 170 and continue on toward one or more reflectors 172 in thedirection indicated by the arrow 174. This acoustic wave energy isreflected by the reflectors 172 and directed back toward the receivingtransducer 170 in the direction indicated by the arrow 176.

The launch and receiving transducers may be connected to separate dipoleantennas. This may be advantageous in certain applications since thedifferent antennas may receive and radiate energy in differentdirections, and this allows separate signal processing for received andtransmitted RF energy.

In FIG. 8, a single launch transducer (LT) 90 transmits surface acousticwaves in both directions to tap transducers (T) 92, 94, 96 and 98. Asmay be seen, the launch transducer 90 is slightly offset (to the left asillustrated in FIG. 8) so that the length of the transmission path 1 tothe tap transducer 92 is shorter than the path 2 to the tap transducer94. Similarly, the path 3 to the tap transducer 96 is shorter than thepath 4 to the tap transducer 98. In particular, the various transducersare positioned such that the differences in propagation times betweenthe pathways 1 and 2, 2 and 3, and 3 and 4 are all equal (ΔT). Theoutputs of the tap transducers 92, 94, 96 and 98 may thus be summed toproduce a second signal S2 of the type represented in FIG. 5.

FIG. 9 illustrates the same basic configuration as in FIG. 8 except thatthe launch transducer 100 operates also to reconvert the SAW energy intoelectrical energy to form the signal S2. Reflectors 102, 104, 106 and108 serve to reflect the acoustic wave energy proceeding on paths 1, 2,3 and 4, respectively, back toward the transducer 100. As in theconfiguration of FIG. 8, the differences in propagation times betweensuccessive pathways (i.e., between pathways 1 and 2, 2 and 3, and 3 and4) are all equal (ΔT).

In the embodiments of FIG. 8 and FIG. 9, surface acoustic wavestraveling along pathways 3 and 4 must pass beneath transducers 92, 94(FIG. 8) or reflectors 102, 104 (FIG. 9). Such an arrangement ofsuccessive, multiple tap transducers or reflectors in a pathwayintroduces unwanted reflections and spurious signals into the outputsignal S2, making subsequent signal processing more difficult.

FIGS. 10 and 11 illustrate SAW device configurations, corresponding toFIGS. 8 and 9, respectively, in which plural launch transducerssimultaneously receive and convert the signal S1 into SAW energy. Withthis arrangement the pathways 1, 2, 3 and 4 are spatially separated sothat the surface acoustic waves can travel on the surface of thesubstrate without passing beneath a reflector or transducer.

It is, of course, possible to combine the configurations of FIGS. 8-11in various ways. FIG. 12 shows an embodiment which combines theprinciples illustrated in FIGS. 9 and 11. In this embodiment, twolaunch/receive transducers 110 and 112 simultaneously convert theinterrogation signal S1 into surface acoustic waves which travel alongpathways 1, 2, 3, 4, 5, 6, 7 and 8. The transducers 110 and 112 arepositioned so that the propagation times along these pathways arestaggered, from one pathway to the next, by a fixed amount ΔT; that is,the propagation time along pathway 2 is ΔT longer than along pathway 1,the propagation time along pathway 3 is ΔT longer than along pathway 2,etc.

It will be appreciated that an information code can be imparted to thesecond (reply) signal S2 by means of “delay pads” of the typeillustrated in FIGS. 6 and 7. These delay pads may be inserted atappropriate places along the respective propagation pathways illustratedin FIGS. 8-12.

The embodiment of FIG. 13 comprises a substrate 120 of piezoelectricmaterial, such as lithium niobate, on which is deposited a pattern ofmetallization essentially as shown. The metallization includes two busbars 122 and 124 for the transmission of electrical energy to fourlaunch transducers 126, 128, 130 and 132. These launch transducers arestaggered, with respect to each other, with their leading edgesseparated by distances X, Y and Z, respectively, as shown. The distancesX and Z are identical; however, the distance Y is larger than X and Zfor reasons which will become clear below.

Further metallization includes four parallel rows of delay pads 134,136, 138 and 140 and four parallel rows of reflectors 142, 144, 146 and148. The two rows of reflectors 144 and 146 which are closest to thetransducers are called the “front rows” whereas the more distant rows142 and 148 are called the “back rows” of the transponder.

The bus bars 122 and 124 include contact pads 150 and 152, respectively,to which are connected the associated poles 154 and 156 of a dipoleantenna These two poles are connected to the contact pads by contactelements or wires 158 and 160, represented in dashed lines.

The embodiment of FIG. 13 is similar, in principle, to the embodiment ofFIG. 12. The provision of four transducers 126, 128, 130 and 132 and tworows of reflectors 142, 144, 146, and 148 on each side of thetransducers results in a total of sixteen SAW pathways of differentlengths and, therefore, sixteen “taps”. These sixteen pathways (taps)are numbered 0, 1, 2 . . . D, E, F, as indicated by the reference number(letter) associated with the individual reflectors. Thus, pathway 0extends from transducer 126 to reflector 0 and back again to transducer126 as shown in FIG. 9. Pathway 1 extends from transducer 128 toreflector 1 and back again to transducer 128. The spatial difference inlength between pathway 0 and pathway 1 is twice the distance X (theoffset distance between transducers 126 and 128). This results in atemporal difference of ΔT in the propagation time of surface acousticwaves.

Similarly, pathway 2 extends from transducer 126 to reflector 2 and backagain to transducer 126. Pathway 3 extends from transducer 128 toreflector 3 and back to transducer 128. The distance X is chosen suchthat the temporal differences in the length of the pathway 2 withrespect to that of pathway 1, and the length of the pathway 3 withrespect to that of pathway 2 are also both equal to ΔT.

The remaining pathways 4, 5, 6, 7 . . . E, D, F are defined by thedistances from the respective transducers launching the surface acousticwaves to the associated reflectors and back again. The distance Y isequal to substantially three times the distance X so that thedifferences in propagation times between pathway 3 and pathway 4 on oneside of the device, and pathway B and pathway C on the opposite side areboth equal to ΔT. With one exception, all of the temporal differences,from one pathway to the next successive pathway are equal to the sameΔT. The SAW device is dimensioned so that ΔT nominally equals 100nanoseconds.

In order to avoid the possibility that multiple back and forthpropagations along a shorter pathway (one of the pathways on the leftside of the SAW device as seen in FIG. 13) appear as a single back andforth propagation along a longer pathway (on the right side of thedevice), the difference in propagation times along pathways 7 and 8 ismade nominally equal to 150 nanoseconds. Specifically, the nominalperiods of propagation (tap delays) along each of the sixteen pathways,and the third signal (mixer) difference frequency resulting from each ofthese tap delays, are as follows:

Pathway (Tap) Tap Delay (nS) Corresponding Frequency 0 900 1000 1 10001281 2 1100 1407 3 1200 1533 4 1300 1659 5 1400 1785 6 1500 1911 7 16002037 8 1750 2232 9 1850 2358 A 1950 2484 B 2050 2610 C 2150 2736 D 22502862 E 2350 2988 F 2450 3114

As shown in FIG. 11, a wavefront produced by reflections from theleading and trailing edges of these fingers will be formed by thesuperposition of a first wave reflected from the first leading edge andsuccessive waves reflected from successive edges and having differencesin phase, with respect to the first wave, of −λ/4, λ/2, −3λ/4, λ, etc.As may be seen, the wave components having a phase −λ/4, λ/2 and −3λ/4effect a cancellation, or at least an attenuation of the wave componentreflected from the leading edge.

The interdigital fingers of the transducers may be split in the mannerillustrated in FIG. 11 to reduce reflections.

Conventional interdigital finger transducers of the type shown in FIG.10, which are constructed to operate at a fundamental, resonantfrequency of 915 MHz, have a finger width (λ/4) of approximately 1micron: a size which approaches the resolution limit of certainphotolithographic fabrication techniques (the selective removal ofmetallization by (1) exposure of photoresist through a mask and (2)subsequent etching of the metallized surface to selectively remove themetal between and outside the transducer fingers). If the fingers aresplit, as shown in FIG. 11, the width of each finger (λ/8) for afundamental frequency of 915 MHz would be approximately ½ micron. Thesize would require sophisticated photolithographic fabricationtechniques.

In order to increase the feature sizes, the transducers in thetransponder are constructed with a resonant frequency f(0) of 305 MHz.In this case, the width of each finger is three times larger thantransducer fingers designed to operate at 915 MHz, so that the width(λ/8) of the split fingers shown in FIG. 11 is approximately 1½ microns.This is well within the capability of typical photolithographicfabrication techniques.

Although the transducers are constructed with a resonant frequency of305 MHz, they are nevertheless driven at the interrogation frequency ofapproximately 915 MHz; i.e., a frequency 3f(0) which is the thirdharmonic of 305 MHz.

FIG. 12 is a diagram showing the energy converted (electrical energy toSAW energy and vice versa) by the transducers employed in the passiveSAW transponder of FIGS. 8 and 9. As may be seen, the transduced energyhas a sin x/x characteristic 170 centered about the fundamentalfrequency f₀ and the odd harmonics: 3f₀, 5f₀, etc. The peak of the sinx/x function falls off abruptly with higher harmonics as indicated bythe dashed line 172. The transducers do not couple any energy into orout of the piezoelectric crystal at the even harmonics (2f₀, 4f₀ . . .).

The energy converted by a transducer, when driven in its third harmonic3f₀ (915 MHz), is about ⅓ of the energy that would be converted if thetransducer were driven at its fundamental frequency f₀ (305 MHz).Accordingly, it is necessary to construct the transducers to be asefficient as possible within the constraints imposed by the system. Asis well known, it is possible to increase the percentage of energyconverted, from electrical energy to SAW energy and vice versa, byincreasing the number of fingers in a transducer. In particular, theconverted signal amplitude is increased by about 2% for each pair oftransducer fingers (either conventional fingers or split fingers) sothat, for 20 finger pairs for example, the amplitude of the convertedsignal will be about 40% of the original signal amplitude. Such anamplitude percentage would be equivalent to an energy conversion ofabout 16%. In other words, the energy converted will be about 8 db downfrom the supplied energy.

The addition of finger pairs to the transducers therefore advantageouslyincreases the energy coupling between electrical energy and SAW energy.However, as explained above in connection with FIGS. 1 and 3, the systemaccording to the invention operates to excite the transducers over arange of frequencies between 905 MHz and 925 MHz. This requires thetransducers to operate over a 20-25 MHz bandwidth: a requirement whichimposes a constraint upon the number of transducer finger pairs becausethe bandwidth of a transducer is inversely proportional to its physicalwidth. This relationship arises from the fact that the bandwidth isproportional to 1/τ, where τ is the SAW propagation time from one sideof the transducer to the other (the delay time across the transducer).

For a transducer driven at its third harmonic of 915 MHz, the spacingbetween successive fingers (single fingers or split fingers) is 3λ2,where λ is the SAW wavelength at 915 MHz. Such a transducer willtherefore be three times as wide as a transducer having the same numberof finger pairs but which is designed with a finger pair spacing of λ/2for a fundamental of 915 MHz.

Therefore, a transducer driven at its third harmonic will have ⅓ thenumber of finger pairs for a given bandwidth than a transducer, alsodriven at that same frequency, which operates at its fundamental orresonant frequency.

FIG. 13 illustrates the transducer structure according to the inventionwhich is designed to maximize energy coupling while providing afrequency bandwidth of approximately 25 MHz at around 915 MHz. Thisdiagram is representational, insofar as the individual fingers of thetransducer are drawn as single lines. It will be understood, however,that these fingers are formed of metallization with a prescribed widthas illustrated in FIG. 11.

The entire transducer is constructed with split fingers as shown in thelefthand section “A” of FIG. 13. For clarity, however, the split fingersin the righthand section “B” have been represented as single lines. Asindicated above, each half finger in the transducer of FIG. 13 has awidth of λ₀/8, where λ₀ is the SAW wavelength at the transducerfundamental frequency of 305 MHz. Stated another way, the width of eachhalf finger in a split finger pair is 3λ/8, where λ is the wavelength ofthe third harmonic at 915 MHz

As shown at the top of FIG. 13, the transducer is divided into severalseparate sections: a central section 180, two flanking sections 182 andtwo outer sections 184. The central section 180 is comprised ofinterdigital transducer fingers which are alternately connected to twoouter bus bars 186 and to a central electrical conductor 188. Thiscentral section comprises a sufficient number of finger pairs to converta substantial percentage of electrical energy into SAW energy and viceversa. By way of example and not limitation, there may be 12 fingerpairs so that the converted amplitude is approximately 24% of theincoming signal amplitude.

Flanking the central section, on both sides, are sections 182 containing“dummy” fingers; that is, fingers which are connected to one electrodeonly and therefore serve neither as transducers nor reflectors. Thepurpose of these fingers is to increase the width of the transducer sothat the outer sections 184 will be spaced a prescribed distance, or SAWdelay time, from the central section 180. For example, there may be 7dummy fingers (or, more particularly, split fingers) in each of thesections 182.

Finally, each of the outer sections 184 of the transducer contains asingle transducer finger pair which is used to shape the bandwidth ofthe transducer of FIG. 13. Referring to FIG. 22 illustrating thecoupling response of the transducer of FIG. 13, there is shown the usualsin x/x function 190 that corresponds to the center section 180 and twosin x/x functions 194 that correspond to the outer transducer sections184. When superimposed, the functions 190 and 194 produce a combinedfrequency characteristic 200 having a widened, and substantially flatupper portion 202 and steeply sloping sides 204. As a consequence, thebandwidth of the transducer of FIG. 13 is increased to the required 25MHz.

According to still another feature of the present invention, thetransducer of FIG. 13 is constructed to closely match the impedance ofthe dipole antenna 154, 156 to which it is connected. This impedancematch maximizes the transfer of energy between the radiation transmittedto and from the antenna and the acoustic energy within the SAW device.

The impedance of a SAW transducer comprises a relatively largecapacitance created by its interdigital fingers plus a small ohmicresistance. With a conventional transducer, this capacitance is severaltimes greater than the reactive impedance (inductance and capacitance)of a microwave dipole antenna designed to operate a 915 MHz. In order toovercome this mismatch, “complex conjugate” matching is employed so thatthe reactive components in the impedance of both the antenna and thetransducers substantially cancel each other. This is accomplished byconstructing each transducer as two serially connected partialtransducers.

FIG. 22 shows two partial transducers 210 and 212 connected in series.If we assume that a conventional transducer having interdigital fingersof total width length L has a capacitance of C₀, then the capacitance ofeach of the transducers 210 and 212, which have total finger widths ofL/2, will be C₀/2. The capacitance, X, of the series connectedtransducer circuit is therefore:2/C ₀+2/C ₀=1/XX=C ₀/4By dividing a conventional transducer into two, serially connectedpartial transducers, the capacitance of the circuit is reduced by ¼.

FIG. 23 illustrates how the two partial transducers 210 and 212 arecombined, in the manner also shown in FIG. 13, to form a singletransducer.

Since, according to the invention, the reactive components of theantenna 154, 156 and the four, parallel connected transducers 126, 128,130 and 132 are closely matched, the power loss within the transponderis limited to the non-reactive (i.e., heat) losses within the antenna,the four transducers and the two bus bars 122 and 124. In this case, thepower coupled into the SAW device is directly proportional to the totalresistance according to the formula:P=r _(total) i ²where the current i remains constant from a “complex conjugate” matchedcircuit and r_(total) is given by:r _(total) =r _(Ω) +r _(a)r_(Ω) in this formula is the total ohmic electrical resistance in theantenna, bus bars and transducers and r_(a) is the equivalent resistance(heat loss) due to the energy coupled into the SAW device substrate.

In order to maximize the energy coupled into the SAW device, it isdesirable to decrease the ohmic resistance rΩ so as to reduce the lossratio:r _(s) /r _(t)=1/(r _(Ω) /r _(a)+1)Particularly when r_(Ω) is approximately the same order of magnitude asr_(a) so that the loss ratio is as low as ½, the value of r_(Ω) is asignificant factor in the power response of the transponder.

Therefore, the bus bars 122 and 124 are made considerably thicker thanthe other metallized elements on the SAW device substrate in order toreduce their ohmic resistance. These bus bars are also made as wide aspossible for the same reason, although the width is not nearly ascritical as the thickness in determining resistance.

The thickness of the bus bars is limited by the effect thatnon-uniformities in thickness may have on the respective phases of thesurface acoustic waves which travel back and forth beneath these busbars. As the bus bar thickness increases, it becomes more difficult tomaintain uniformity so that variations in phase, due to differences inmass loading between the various paths of travel, may occur.

The bus bars 122 and 124 are made approximately twice as thick as theother metallized elements on the substrate, as represented in thecross-sectional view of FIG. 16. Specifically, the transducers, delaypads and reflectors are formed by depositing 1000 Angstroms of aluminumon the substrate, whereas the bus bars are formed by depositing 2000Angstroms of aluminum over 300 Angstroms of chromium. The chromiumprovides a good bond between the aluminum and the lithium niobatesurface.

In practice, the metallization is deposited on the substrate surfaceusing a two-layer photolithographic process. Two separate reticles areused in forming the photolithographic image: one reticle for thetransducers, reflectors and phase pads as well as the alignment marks onthe substrate, and a separate reticle for the bus bars.

FIG. 25 illustrates the equivalent circuit of the transponder. Shown atthe left of the diagram are the capacitance C_(a) and inductance L_(a)of the dipole antenna. This reactive impedance is matched, as closely aspossible, to the reactive impedance C₀ of the four transducers so thatonly the resistance losses r_(Ω) and r_(a) affect the coupling. Also theohmic resistance r_(Ω) is reduced as much as possible by increasing thethickness and width of the bus bars.

FIG. 26 shows, in enlarged scale, a single reflector 220 and three delaypads 222. Each of the delay pads is illustrated as having a centralmetallized portion 224 and two edge portions. The edge portions 226 ofthe delay pads, as well as the lateral edges of the bus bars 122 and 124(i.e., the edges transverse to the SAW paths of travel) are providedwith two levels of serrations to substantially reduce SAW reflectionsfrom these edges.

FIG. 27 illustrates the lower right-hand corner of a delay pad 222 ingreatly enlarged scale. As may be seen, the serrations comprise twosuperimposed “square waves” having the same pulse height but differentpulse periods. The pulse height for both square waves is λ/4. By way ofexample and not limitation, the pulse period is λ/3 for one square waveand 6λ for the other, where λ is the SAW wavelength at 915 MHz

The affect of the serrations in canceling reflections is indicated atthe center of FIG. 19. Shown there are the SAW reflections from thevarious levels of the delay pad edge. It will be understood that thewavefronts of reflections having phases of λ/2 will be equal andopposite in phase to the reflected wavefronts having phases of 0 and λ.

The first level of serrations serves to reduce reflections, while thesecond level serves to break up the average reflection plane.

Unlike the transducers, the reflectors 0, 1, 2 . . . E, F used in theembodiment of FIGS. 8 and 9 are designed for a fundamental frequency of915 MH. Therefore, the distance between successive reflector fingers isλ/2 and the width of each finger is approximately λ/4, where λ is theSAW wavelength at 915 MHz. Each reflector has, for example, a total of20 fingers.

FIG. 14 illustrates a particular reflector 240 which may be employed asreflectors 0, 1, 2, . . . E, F in the embodiment of FIG. 8. Thereflector 240 comprises a plurality of fingers 242 (in this case 20)connected between two shorting bus bars 244 and 246. Each two successivefingers are also shorted at one or more locations between the bus bars.Thus, in the embodiment shown, the two fingers 242 on the right-handside of the reflector are shorted at four locations by interconnectingmetallization 248. The shorts between successive fingers reduce energyloss due to ohmic resistance of the fingers and render the reflectorless susceptible to fabrication errors.

As noted above, the distance between successive fingers in the reflectormust be equal to λ/2, where λ is the SAW wavelength at 915 MHz.Consequently, the width of each finger must be somewhat less than λ/2:for example, approximately λ/4 or about 1 micron. Such a finger widthnudges the lower boundary of conventional photolithographic fabricationtechniques so that one or more fingers of a reflector may, in practice,be interrupted along their length. The short circuit bridges betweensuccessive fingers make it possible to retain the function of allfingers of a reflector, although one or more fingers may not extend theentire distance between the bus bars 244 and 246.

In summary, the present invention provides a number of features in apassive SAW transponder which (1) reduce interference caused by unwantedSAW reflections in the transponder substrate and (2) increase thepercentage of energy coupled into and out of the substrate.

FIG. 15 is a graph illustrating the ranges of amplitudes which areexpected in the individual components of the second (reply) signalassociated with the respective pathways or tap delays 0-F. As may beseen, the greatest signal amplitudes will be received from pathwayshaving reflectors in their front rows; namely, pathways 0, 1, 4, 5, 8,9, C and D. The signals received from the pathways having reflectors intheir back rows are somewhat attenuated due to reflections andinterference by the front row reflectors. If any one of the amplitudesa_(i) at one of the sixteen frequencies f_(i) in the third signal fallsoutside its prescribed range, the decoded identification number for thattransponder is rejected.

As indicated above, transponders of the type illustrated in FIGS. 6-13are susceptible to so-called “manufacturing” variations in response, dueto manufacturing differences from transponder to transponder, as well astemperature variations in response due to variations in ambienttemperature. Particularly the case where small differences in tap delaysin the order of one SAW cycle period are measured to determine theencoded transponder identification number, these manufacturing and/ortemperature variations can be in the order of magnitude of theinformational signal. It is desirable, therefore, to provide systems forcompensating both manufacturing and temperature variations. Such systemswill now be described in connection with the embodiment of FIGS. 1 and13.

As explained above, the transponder identification number contained inthe second (reply) signal is determined by the presence or absence ofdelay pads in the respective SAW pathways. These delay pads make aslight adjustment to the propagation time in each pathway, therebydetermining the phase of the surface acoustic wave at the instant of itsreconversion into electrical energy at the end of its pathway.Accordingly, a fixed code (phase) is imparted to at least two pathwaysin the SAW device, and the propagation times for these pathways are usedas a standard for the propagation times of all other pathways. Likewise,in a reflector-based acoustic device, a reflector may be provided at apredetermined location to produce a reference signal.

The mask variation ΔM_(i) for a given pathway, i.e., a variation in tapdelay due to imperfections in the mask—will be the same for alltransponders made from the same mask.

The time variations ΔO_(i) is the so-called “offset” variation which isprimarily due to variations in the interdigital finger line widths of areflector in the front row through which the surface acoustic waves mustpass to reach a reflector i in the back row. Variations in transducerfinger line widths are already reflected in the initial pathwaypropagation time T₀.

Since the time variations ΔR_(i), are completely random from pathway topathway and from transponder to transponder, it is not possible tocompensate for these. If a random variation ΔR_(i), becomes too large,however, the transponder identification number reading will be rejected,since one of the amplitudes a_(i) or phases Φ_(i) will fall outside ofthe acceptable limits.

It may be seen, there are primarily three types of variations which mustbe compensated:

(1) Variations due to temperature which are reflected in large changesin the propagation times T₀ and ΔT. These temperature variations aresubstantially (but not exactly) the same for each pathway.

(2) Mask variations ΔM which are different for each pathway but the samefor all transponders manufactured from a given mask.

(3) Offset variations ΔO which are primarily due to the effect ofvariations in the line widths of front row reflectors on the pathways toback row reflectors. These variations are traceable to the manufacturingprocess (such as the mask exposure time) and are normally the same forall parallel front row reflectors on one side of a transpondersubstrate. The line widths may vary from one side of the substrate tothe other due to lack of orthogonality in the mask exposure.

The three types of variations identified above—namely, temperature, maskand offset variations—are compensated as follows:

(1) Temperature variations are compensated by determining the times T₀and ΔT from two successive pathways i and j to provide a firsttemperature estimate, and then compensating small, second ordervariations by averaging the propagation times of the four front rowpathway pairs (pathways 0 and 1, 4 and 5, 8 and 9 and C and D).

(2) The variation ΔM, which relates to the mask, will be the same forall transponders made from the same mask. Consequently, this variationmay be isolated and compensated for by determining the amplitudes a_(i)and phases Φ_(i) for a large number of transponders, and thereafterdetermining statistically the acceptable limits for these parameters. Byway of illustration, the amplitudes a_(i) from different transpondersmade from the same mask for each frequency f_(i) may be plotted on agraph such as that shown in FIG. 15 to determine their statisticaldistribution. The acceptable limits of amplitude may then be determinedfor each frequency from this statistical distribution. FIG. 15 shows onesuch distribution curve 170 of amplitudes for the frequency 2.45 kHz(pathway F).

Variations in the phases Φ_(i) of different transponders traceable tothe mask are compensated in a similar manner by adjusting the centerphases (nominally 0°, 90°, 180° and 270°) and the phase tolerances(nominally +/−30° about each center phase) for each “phase bin”.

After the initial compensation for mask variations ΔM, all subsequentmasks used to manufacture transponders may be adjusted so as to matchthe imperfections in the original mask. The mask variations ΔM aretherefore caused to remain identical for all transponders used in agiven system.

(3) Finally, offset variations ΔO, which are traceable to manufacturingprocess variations, are compensated by determining ΔO_(F) and using thisvalue as a standard to eliminate the effect of offset in all the “backrow” pathways; i.e., pathways 2, 3, 6, 7, A, B and E.

The entire process of compensation is illustrated in the flow chart ofFIG. 16. As is indicated there, the first step is to calculate theamplitude a_(i) and phase Φ_(i) for each audio frequency Φ_(i) (block180). Thereafter, the sixteen amplitudes are compared against theiracceptable limits Φ_(i) (block 182). As shown in FIG. 15, these limitsmay be different for each amplitude. If one or more amplitudes falloutside the acceptable limits, the transponder reading is immediatelyrejected.

If the amplitudes are acceptable, the phase differences Φ_(ij) arecalculated (block 184) and the temperature compensation calculation isperformed to determine the best value for ΔT (block 186). Thereafter,the offset compensation calculation is performed (block 188) and thephases for the pathways 2, 3, 6, 7, A, B and E are adjusted.

Finally, an attempt is made to place each of the pre-encoded phases intoone of the four phase bins (block 190). If all such phases fall within abin, the transponder identification number is determined; if not, thetransponder reading is rejected.

Active Transponder First Embodiment

Typical semiconductor memory active tag system are described in U.S.Pat. Nos. 4,739,328; 4,782,345; 5,030,807; 4,999,636; 5,479,160;3,914,762; 5,485,520; 4,123,754; 5,086,389; 5,310,999; 4,864,158;4,853,705; 4,816,839; 5,055,659; 4,835,377; 4,912,471; 4,358,765;4,075,632; and 3,984,835 incorporated herein by reference.

U.S. Pat. No. 4,739,328 provides a system which interrogates an activesemiconductor memory tag, as shown in FIGS. 30 and 39-43. The systemproduces pluralities of cycles of signals at first and second harmonicfrequencies. The cycles of the signals identifying a binary “1” aresymmetrical to the signals identifying binary “0”. In other words, thecycles at the second frequency occur before a cycle at the firstfrequency to represent a binary “1” and the cycle at the first frequencyoccurs before the cycles at the second frequency to identify a binary“0”. A code is also provided, different from a “1” or a “0”, indicatingthe end of the transmission of signal cycles. The system also provides ageneration of clock signals on a self-synchronizing basis regardless ofthe patterns of binary 1's and binary 0's transmitted to the reader.This facilitates the detection by the reader on a straightforward andreliable basis of the sequence of binary 1's and binary 0's identifyingthe object. In other words, the data transmission format supports anefficient clock recovery system.

The reader receives the signal cycles identifying the object and delaysthese signal cycles by (a) a first time such as one fourth (¼) of theperiod of a cycle at the second frequency, (b) a second time such as onehalf (½) of such period and (c) a third time such as such one (1) suchperiod. The reader compares the received signal cycles and the firstdelayed signal cycles to produce first phase-locked signals forgenerating the clock signals. The reader compares the received signalcycles and the second delayed signal cycles to produce additionalphase-locked signals at the times that the first phase-locked signalsfor generating the clock signals are not produced. The reader thenproduces clock signals from the first and additional phase-lockedsignals. The reader also produces information signals from a comparisonof the received signal cycles and the third delayed signal cycles.

A source 410 of interrogating RF signals is connected to an antenna 412at a reader generally indicated at 414. The interrogating RF signalsfrom the source 410 may have a suitable frequency such as 915 MHz. Whenthe source 410 of interrogating RF signals is energized, the antenna 412transmits activating signals to a suitable antenna 416 (such as a dipoleantenna) at a transponder generally indicated at 418. The transponder418 is located at an object (not shown) to identify the object. Thetransponder includes a data source such as a read-only memory 422 whichprovides a sequence of binary 1's and binary 0's in an individualpattern.

A binary “1” in the read-only memory 422 causes a modulator 420 toproduce a first plurality of signal cycles and a binary “0” in theread-only memory 422 causes the modulator 420 to produce: a secondplurality of signal cycles different and distinguishable from the firstplurality of signals. The pluralities of signal cycles sequentiallyproduced by the modulator 420 to represent the pattern of binary 1's andbinary 0's identifying the object are introduced to the dipole 416 fortransmission to the antenna 412 at the reader.

The antenna 412 introduces the received signals to a mixer 426 forcomparison in the mixer with the interrogating RF signals from thesource 410. The mixed signals are introduced to an amplifier 428 and aredemodulated in a demodulator 430. The demodulator produces signals in asequence having a pattern identifying the pattern of 1's and 0's in theread-only memory 422 at the transponder.

A reader, generally indicated at 424, is shown in detail in FIGS. 39 a,39 b, 39 c and 39 d and may be considered to be similar in some detailsto that shown in FIG. 30 and described above. The signals aretransmitted by the dipole 416 in FIG. 30 to an antenna 426 (FIG. 39 a)and are introduced to mixers 432 and 434. The interrogating RF signalsfrom a source 430 (corresponding to the source 410 in FIG. 30) are alsointroduced to the mixer 432 and are shifted in phase by 90° at 436 andare then introduced to the mixer 434. The signals from the mixers 432and 434 are respectively amplified linearly at 440 and 442 and are thenintroduced to a mixer or third channel combiner 444. The signals fromthe amplifiers 440 and 442 and from the mixer 444 are respectivelyintroduced to amplifiers 446, 448 and 450, each of which provides a highgain and then a limitation in amplitude after providing such high gain.

The signals from the limiting amplifiers 446, 448 and 450 respectivelypass to shift registers (FIG. 39 b) 454 and 456 in series, 458 and 460in series and 462 and 464 in series. The signals at mid points in theshift registers 454, 458 and 462 respectively pass to exclusive OR gates466, 468 and 470. Signals are also respectively introduced to the inputterminals of the exclusive OR gates 466, 468 and 470 from the outputterminals of the limiting amplifiers 446, 448 and 450.

Connections are made from the output terminal of the exclusive OR gate466 to input terminals of AND gates 472 and 474, from the outputterminal of the OR gate 468 to input terminals of the AND gate 472 andan AND gate 476 and from the output terminal of the OR gate 470 to inputterminals of the AND gates 474 and 476. The output terminals of the ANDgates 472, 474 and 476 are connected to input terminals of an OR gate480. (FIG. 39 c). The AND gates 472, 474 and 476 are shown in FIGS. 39 band 39 c.

In like manner, the signals from the limiting amplifiers 446, 448 and450 (FIG. 39 b) are respectively introduced to exclusive OR gates 482,484 and 486. Second input terminals of the exclusive OR gates 482, 484and 486 respectively receive signals from the output terminals of theshift registers 454, 458 and 462. Signals are introduced from the outputterminal of the exclusive OR gate 482 to input terminals of AND gates490 and 492, from the output terminal of the exclusive OR gate 484 toinput terminals of the AND gate 490 and an AND gate 494 and from theoutput terminal of the exclusive OR gate 486 to input terminals of theAND gates 492 and 494. The output terminals of the AND gates 490, 492and 494 are connected to input terminals of an OR gate 496. (FIG. 39 c).The AND gates 490, 492 and 494 are shown in FIGS. 39 b and 39 c.

Correspondingly, the output terminals of the shift registers 456, 460and 464 (FIG. 39 b) respectively pass to input terminals of exclusive ORgates 500, 502 and 504. The exclusive OR gates 500, 502 and 504 alsohave input terminals respectively connected to the output terminals ofthe limiting amplifiers 446, 448 and 450. Connections are respectivelymade from the output terminals of the exclusive OR gates 500, 502 and504 to input terminals of AND gates 510 and 512, input terminals of theAND gate 510 and an AND gate 514 and input terminals of the AND gates512 and 514. The signals passing through the AND gates 510, 512 and 514are introduced to input terminals of an OR gate 116. (FIG. 39 c). TheAND gates 510, 512 and 514 are shown in FIGS. 39 b and 39 c.

The signals from the OR gate 480 pass to a phase-locked loop 520 whichmay be constructed in a conventional manner. The phase-locked loop mayinclude a phase detector and a voltage-controlled oscillator. An outputterminal of the phase detector of the phase-locked loop 520 is connectedthrough a switch 522, controlled by the signal from the OR gate 496, toselectively provide feedback to the voltage controlled oscillator. Thesignals from the voltage-controlled oscillator of the phase-locked loop520 are introduced to the clock terminal of a low-pass digital filter532.

When the dipole 416 at the transponder 418 in FIG. 30 receives theinterrogating RF signals from the reader 424, it generates pluralitiesof signal cycles in a code dependent upon the pattern of 1's and 0'sprovided in the data source such as the read-only memory 422 to identifythe object associated with the transponder. For example, a binary “0”may be represented by a first signal cycle at a suitable frequency suchas twenty kilohertz (20 kHz) and then by additional signal cycles at asuitable frequency which is a harmonic of the first frequency. Forexample, the second frequency is forty kilohertz (40 kHz) when the firstfrequency is twenty kilohertz (20 kHz). This is illustrated at 540 inFIG. 40. Similarly, a binary “1” may be represented by two (2) cycles atthe second frequency such as forty kilohertz (40 kHz) and then by anadditional cycle at the first frequency such as twenty kilohertz (20kHz). This is illustrated at 542 in FIG. 40.

The read-only memory 422 causes the modulator 420 to produce thepluralities of signal cycles respectively coding for the sequences ofbinary 1's and binary 0's in the individual pattern. The modulator 420introduces these signals to the dipole 416 (FIG. 30) for transmission tothe reader 424. These signals are received by the antenna 426 (FIG. 39a) and are introduced to the mixers 432 and 434 (FIG. 39 a). The mixer432 also receives the interrogating RF signals from the source 430 andthe mixer 434 receives the interrogating RF signals from the source 430after the signals have been shifted in phase by 90° by the stage 436.The mixed signals from the mixers 432 and 434 respectively pass throughthe amplifiers 440 and 442 to the limiting amplifiers 446 and 450. Thesignals from the amplifiers 440 and 442 are also mixed in the stage 444and these mixed signals are introduced to the limiting amplifier 448.

If only one mixer such as the mixer 432 were used, the output from themixer could disappear or become null if the received signal happened tobe in quadrature phase (90° or 270°) with respect to the interrogatingRF signal. By providing the mixers 432 and 434 and by providing an 90°shift in phase in the interrogating signal introduced to the mixer 434,a null cannot simultaneously occur at both of the mixers. As a result,an output signal will pass from at least one of the mixers under all ofthe different phase relationships possible between the received signaland the interrogating RF signal.

There is still one possibility of a null in the output. This may occurwhen the outputs of the mixers 432 and 434 have opposite polarities.That is, the output of one of the mixers 432 and 434 may be the inverseof the output of the other mixer. To prevent a null from occurring undersuch circumstances, the combiner 444 is included to combine the outputsof the signals from the linear amplifiers 440 and 442. The signals fromthe combiner 444 are introduced to the limiting amplifier 448. Sinceoutputs are obtained from the three limiting amplifiers and since theoutputs of these amplifiers are paired (e.g., in the AND gates 472, 474and 476), an output is obtained from at least two (2) of these three (3)AND networks under all possible circumstances.

The shift registers 454 and 456 (FIG. 39 b) delay the signals from theamplifier 546. The delay provided by the shift register 454 correspondsto one-half of the period of a signal cycle at the second frequency suchas forty kilohertz (40 kHz). This signal cycle is introduced to theexclusive OR network 482. (FIG. 29 b). Similarly, the exclusive ORnetwork 466 receives from the shift register 454 the signal cyclesdelayed by a quarter of the period of a signal at the second frequencysuch as forty kilohertz (40 kHz). The shift register 456 provides thesame delay as the shift register 454 so that the signal cycle introducedto the exclusive OR network 500 has a phase shift corresponding to acomplete period of a signal cycle at the second frequency such as fortykilohertz (40 kHz). The exclusive OR networks 468, 484 and 502 receivesignal cycles respectively delayed by the same time period as the signalcycles received by the exclusive OR networks 466, 482 and 500. This samephase relationship is also present in the signal cycles introduced tothe exclusive OR networks 470, 486 and 504.

The exclusive OR gate 466 compares the amplitude of the signal cyclesfrom the amplifier 446 with the amplitude of the delayed signal cyclesfrom the shift register 454. When the amplitudes of the signal cyclesare both simultaneously high or are both simultaneously low, theexclusive OR gate 466 produces a signal with a high amplitude. At allother times, the signals from the exclusive OR gate 466 have a lowamplitude. The exclusive OR gates 468 and 470 respectively provide asimilar comparison of the signal cycles from the amplifier 448 and thedelayed signals from the shift register 458 and with the signal cyclesfrom the amplifier 450 and the delayed signals from the shift register462. The signals passing through the exclusive OR gates 466, 468 and 470are introduced in individually paired relationships to the AND gates472, 474 and 476 (FIGS. 39 b and 39 c). The AND gates 472, 474 and 476in turn pass signals to the OR gate 480 which operates to provide anoutput signal when it simultaneously receives signals of high amplitudesfrom two of the AND gates 472,474 and 476.

In effect, the exclusive OR gates 466, 468 and 470 and the AND gates472, 474, 476 and 480 operate to provide a comparison of the amplitudesof the received signal cycles and the received signal cycles delayed byone quarter of a time period of a signal cycle at the second frequencysuch as forty kilohertz (40 kHz). This comparison is indicated in FIG.42. In FIG. 42, the received signal cycles are indicated at 546 and thedelayed signal cycles are indicated at 548. The output from the OR gate80 is indicated at 550 in FIG. 42. This output has a frequency whichconstitutes the fourth harmonic (e.g. 80 KHz) of the first frequencysuch as twenty kilohertz (20 kHz). The signals 550 occur in most cyclesat the third frequency (e.g. 80 kHz) but, as will be seen at 552, do notoccur in all cycles. The signals 550 have a high amplitude when theamplitudes of the signal cycles 546 and 548 are simultaneously high orsimultaneously low.

The exclusive OR gates 482, 484 and 486 respectively compare theamplitudes of the signals from the limiting amplifiers 446, 448 and 450with the amplitudes of the output signals from the shift registers 454,458 and 462. This comparison is indicated in FIG. 43. As will be seen inFIG. 43, the signal cycles from the amplifiers 446, 458 and 450 areindicated at 546 and the signal cycles from the phase shifters 454, 458and 462 are indicated at 554. As a result of the comparison, signals areproduced as indicated at 556 in FIG. 43. The signals 556 have a highamplitude when both the signals 544 and 554 simultaneously have a lowamplitude or simultaneously have a high amplitude. The signals 556 areproduced at the third frequency (e.g. 80 kHz) at the times that thesignals 550 in FIG. 43 are not produced at this frequency. As a result,when the signals 550 and 556 are combined, the combination occurs at aperiodic rate corresponding to the third frequency such as eightykilohertz (80 kHz).

The signals 550 passing through the OR gate 80 (FIG. 39 c) areintroduced to the phase detector of the phase-locked loop 520 in FIG. 39c to obtain the production by the oscillator of signals at a particularfrequency. e.g., 1.28 MHz. which is a harmonic of both the firstfrequency of twenty kilohertz (20 Hz) and the second frequency of fortykilohertz (40 kHz) and is also a harmonic of the phase-locked signals atthe third frequency of eighty kilohertz (80 kHz) from the OR gate 480and 496. As a result, the signals from the OR gate 480 and 496constitute phase-locked signals to obtain the generation by thevoltage-controlled oscillator in the phase-locked loop 520 of the clocksignals at the frequency of 1.28 MHz.

The clock signals at the frequency of 1.28 MHz from thevoltage-controlled oscillator in the phase-locked loop 520 areintroduced to the low pass digital filter 532. The filter 532 alsoreceives the signals passing through the OR gate 516. The operation ofthe OR gate 516 may be seen from FIG. 41. In FIG. 41 the received signalcycles are indicated at 546 and the delayed signal cycles (delayed byone full cycle at 40 KHz); from the shift registers 456, 460 and 464 areindicated at 568. The results of the comparison between the signalcycles are indicated at 570. The signal cycles 570 represent thedemodulated signals identifying the object associated with thetransponder 418. The filter 532 filters the demodulated signals 570 topass only the low frequencies represented by the demodulated signals andto prevent the passage of short pulses representing noise.

FIG. 39 d illustrates on a somewhat simplified schematic basis a systemfor utilizing a sequence of signals 578 (FIG. 40) coding for the end ofthe sequence of binary 1's and binary 0's identifying the objectassociated with the transponder 418 and also coding for the beginning ofthe next such sequence. The system shown in FIG. 39 d includes thevoltage-controlled oscillator in the phase-locked loop 520 and afrequency divider 580. The frequency divider 580 receives the signalsfrom the voltage-controlled oscillator 520 and divides these signals toproduce clock signals at the second frequency such as forty kilohertz(40 kHz). These signals are introduced to a terminal of a shift register586, another input terminal of which is connected to receive thedemodulated signals 570. (FIG. 41).

The shift register 586 has six output terminals each of which isconnected to the shift register to produce an output upon a successiveoccurrence of one of the signals from the frequency divider 580. Whenthe six output terminals from the shift register 586 simultaneously havesignals of high amplitude, a signal passes through an AND gate 588. Thissignal indicates that the transmission of the pluralities of signalcycles from the transponder 418 has been completed and that a newsequence of such transmission is being initiated. The signal from theAND gate 588 is introduced to an AND gate 590, another terminal of theAND gate being connected to receive the demodulated signals 570. Theoutput from the AND gate 590 accordingly synchronizes the start of a newtransmission of the pluralities of signal cycles identifying the object.

U.S. Pat. No. 4,888,591, incorporated herein by reference, and explainedwith respect to FIGS. 31-38, discloses the use of the active tag systemwith a spread spectrum interrogation signal, in order to obtain spatialselectivity.

The spread spectrum transmitter includes a 915 MHz oscillator whichprovides the carrier signal. The carrier signal from oscillator ispassed to phase modulator to modulate it with a modulating signal havingcharacteristics which provide unity output when correlated with itselfwith zero time shift, and a substantially lower output level whencorrelated with itself with significant time shift. For example, randomand pseudo-random modulating signals have these characteristics. Such asignal could be generated, for example, by a stationary stochasticprocess. In this embodiment, the modulating signal is binary, employinga pseudo-random pulse sequence. As will be described in more detaillater, this pulse sequence is supplied by sequence a generator throughdriver-shaper. Driver-shaper rounds the edges of the pulses to improvetheir shape for easier modulation onto the carrier. The sequencegenerator is timed by a shift clock generator.

The modulated output signal from phase modulator is sometimes referredto as a direct spread spectrum signal. The phase modulator produces adouble-sideband, suppressed carrier signal which is passed to bandpassfilter, which limits the spectrum to that permitted by regulation. Anamplifier amplifies the signal to be transmitted to raise it to thedesired power level needed for transmission.

Where the modulation signal produced by the sequence generator is apseudo-random signal, a shift register type generator may be employed.The pseudo-random output code sequence is connected to a driver/shaper.Three register stages provides a sufficiently high repeat cyclefrequency, repeating every seven clock cycles. If clocked at a rate of10 MHz, this frequency is usually sufficiently high so as to be free ofconflict with the backscatter-modulated return signals, which usuallyhave a much lower frequency, for example, 20 and 40 kHz. In this system,it is important to bear in mind that there is only a slight delayintroduced by the transponder, with the information code generated by amodulator.

Using additional shift register stages, which produce a longer repeatsequence of the pseudo-random modulation signal, provides even bettercorrelation differentiation. This means that the amplitude differencesbetween correlated return signals, which are closer to the reference,and less correlated ones, which are usually the ones to be eliminated,are greater, potentially enabling distinction-of unwanted signals.Usable shift register sequences are described in Shift RegisterSequences, Solomon W. Golomb, Holden-Day, Inc. (1957), Ch. 3.

As disclosed in U.S. Pat. No. 4,888,951, a conventional, high frequencycirculator is provided, which passes signals only in one direction, fromone of ports to the next. Thus signals from a directional coupler arethus passed out onto an antenna. Backscatter-modulated signals receivedby antenna from the tag are passed into the circulator and are thencirculated to a mixer. The mixer compares a reference signal from thedirectional coupler with the returned signal from the antenna. The mixeris, for example, a double-balanced mixer used as a phase comparator. Themixer compares the phases (typically 0 degrees or 180 degrees) of itstwo input signals. The output signal from mixer is passed through alowpass filter to a preamplifier, whose output then is ported to areceiver/detector. The mixer is switched by a reference signal. When thereference signal is one polarity, the modulated backscattered signalpasses directly through the mixer. When the reference signal is of theopposite polarity, the modulated backscattered signal is inverted. Thelowpass filter serves to average the amplitude of the output signalsfrom the mixer. This averaging results in a signal having a relativelylow amplitude where the reference and the returned signals are out ofmacro phase, and a high amplitude where they are in macro phase of thephase modulation signal.

The instantaneous amplitude of the IF output signal from the mixerdepends primarily upon two factors: (1) the amplitude of the RF inputsignal; and (2) the cosine of the relative phase angle between thereference signal and the RF signal.

A pulse sequence is the sequence, generated by a sequence generator,pulse shaped, and passed to a phase modulator and through a bandpassfilter. The signal sequence is then amplified and transmitted throughthe antenna to the tag, where it is backscatter-modulated by an activeelement with an information signal sequence generated in the tag andreturned through the antenna to the input of circulator.

Two kinds of phase changes occur to the modulated backscattered signal.There is a micro-phase change which is the shift in the modulatedbackscattered signal, containing the necessary data. These micro-changesare an undesired side effect in signal decoding, and are readily dealtwith in systems of the prior art such as those described in U.S. Pat.No. 4,739,328. There is also a macro-phase change wherein the entireenvelope of the signal shifts its phase.

The general appearance of the returned signal remains as shown inwaveform 372 in FIG. 35, but the amplitude will change slightly inaccordance with the backscatter modulation, and the overall macro-phase(the envelope) will shift during the transmission to and from the tag.This phase relationship can best be illustrated using square waves.Comparing the actual reference signal and the illustrative square wavesignal 370 shows that the dots at the axis of signal 372 correspond tomacro-shifts in phase from high-to-low and low-to-high of the referencesignal 370, corresponding to 180 degree phase shifts of the RF signal.For the purposes of illustration, it is assumed that during signaltransmission to the tag and return, the frequency of the referencesignal was shifted by the width of one pulse of the modulating signal.

The operation of mixer 352 is such that if the reference sample on line351 and the RF sample online 358 are perfectly in phase-modulatingphase, the maximum output signal amplitude from mixer 352 on line 360will be obtained. Thus if the reference and returned RF signals are bothpositive, the output signal is positive; if they are both negative, theoutput signal is also positive; and if one is positive and the othernegative, the output signal is negative.

For illustration, let us assume that the returned RF signal 373 isshifted in phase from the reference signal, as shown, by one completepulse width. The pulses of reference signal 370 have been numbered from1 to 7. Similarly, the pulses of the returned RF signal 373 have beennumbered from 1 to 7. Looking at pulse 1 of returned signal 323, whichis a HIGH, and comparing it with the pulse 2, the coincident pulse intime of the reference signal 370, both pulses are high, providing acorrelation of +1, shown beneath pulse 1 of waveform 373. Pulse 2 ofwaveform 373 also correlates exactly with coincident pulse 3 of thereference stream 370, and a correlation of +1 is shown below for thatpulse as well. Pulses 3, 4 and 5 of the returned stream do notcorrelate, each resulting in correlations of −1, as shown. Pulse 6 of RFreturn signal 373 does correlate with pulse 374 of the reference signal,producing a correlation of 1, but pulse 7 does not, again producing acorrelation of −1. Adding the correlation factors of the seven pulses ofthe returned signal 373, the sum is −1. Since there were seven pulses,the averaged correlation is − 1/7.

Next, assume that the returned signal was shifted by only one-half pulsewidth rather than a whole one, as illustrated by returned signal 374. Itis clear that the first two and one-half pulses of returned RF signal374 correlate with the coincident pulses of reference signal 370, asshown, providing a correlation factor of +2½. The next one-half pulse ofreturned stream 374 does not correlate, producing a −½ correlation. Thenext one-half pulse does correlate; the next one-half does not; the nextone-half does; and the next one-half does not, netting for the total ofthe two full pulse durations a correlation factor of zero. The next oneand one-half pulses correspond, for a correlation factor of +1½, and thelast one-half pulse does not correspond, for a −½. The sum of thesecorrelation factors is 3 which, averaged for 7 pulses, is an averagedcorrelation of 3/7.

Referring to FIGS. 34 and 35, the correlation between the returned RFsignal on line 358 and the reference signal on line 351 controls theaveraged amplitude of the signal on line 359 from mixer 352. It will beapparent that a returned signal which correlates as poorly as signal 373does to reference signal 370 will produce an output signal from mixer352 having an averaged amplitude 1/7 the averaged amplitude of a signalwhich correlates 100%. Returned signal 374, which was shifted only byone-half pulse width from reference signal 370, produced an averagedcorrelation of 3/7, meaning that the output amplitude of that returnedsignal from mixer 352 on line 359 would be about half the amplitude of afully correlated signal. Similarly, the output amplitude of signal 373,shifted from the reference signal 370 by a full pulse width, has a levelonly 1/7 of that of a fully in-phase returned signal.

The output signal from mixer 352 on line 360 is passed to a preamplifier361, producing a preamplified signal at node B shown in FIG. 34 whoseamplitude is dependent on the phase correlation in mixer 352.

Referring now to FIG. 31, the demodulated signal at node B passes todecoder/demodulator 312. Decoder/demodulator 312 demodulates and decodesthe signal to extract the binary code which was modulated onto thereturned, modulated backscattered signal by tag 313.

As a result of the operation of receiver/detector 311 described above,the amplitude of the output signal passing through node B todecoder/demodulator 312 varies significantly depending upon the degreeof correlation between the phase modulation pattern on the referencesignal from transmitter 310 and the modulated backscattered signal fromthe tag 313. As explained above in connection with the graphs in FIG.35, the amplitude of this output signal is substantially lower, by afactor of two or more, if the phase correlation between these twosignals is poor, and substantially higher if the correlation is good.Where the correlation is poor, the input signal at node B todecoder/demodulator 312 is of a sufficiently low amplitude below thedecoder/demodulator sensitivity, and is thus ignored. Where thecorrelation is good, the amplitude is sufficiently high so thatdecoder/demodulator 312 can decode the signal and reproduce the originalbinary bit pattern modulated onto the returned signal by the tag.

The graphs of FIGS. 36 and 37 illustrate the advantages of the DSSSsystem. The first portion of the curve on FIG. 37 for a distance between5 and 25 feet shows the usual falloff of signal strength obtained with asystem of the prior art without using the spread spectrum signalmodulation according to the invention. The curve has been normalized toshow a maximum signal strength of 1.0 at 5 feet from the antenna Note inthis first portion of the curve in FIG. 37 that the falloff in signalstrength at 15 feet is about 70%, and the falloff at 25 feet is almostcomplete. Accordingly, using a stationary frequency system, it would bevery difficult to discriminate, based upon this small amplitudedifferential in signal (30%), between a proper signal generated 15 feetaway from the antenna and a spurious one from 25 feet away.

The graph of FIG. 37 ignores the normal signal decay with distance shownin FIG. 36, and only takes into consideration the falloff in signalstrength with distance based upon the modulation technique used in thesubject invention. At 25 feet, the lack of correlation shown by thesignificant drop in signal level results from the DSSS technique. Thesignal strength falls almost to 0. It turns out that there is a nullpoint at 25 feet, but even well beyond 25 feet, up to about 1,500 feetfor a 915 Mhz carrier, the signal strength increases only a smallamount, well below 10%. At about 37 feet, for example, the signal levelis 1/7, or 0.15, as described earlier and shown in FIG. 37, due to lackof correlation. Accordingly, it is very easy to discriminate between adesired signal 15 feet from the reader, and an unwanted signal, such asfrom an adjacent toll lane, which in most cases will be at least 25 feetaway.

It is possible to tailor the distances in actual set up very accuratelyby locating the antenna at the desired distance from the tag even thoughthe transmitter, receiver/detector and decoder are located somewhereelse. They may be connected by shielded cable. The antenna is the onlycomponent of the system whose location is critical.

The operation of the apparatus in toll lanes is illustrated in FIG. 38.The apparatus 370 has its antenna 371 located adjacent LANE 1 of thetoll lane, as shown. No matter where the automobile being monitored inLANE 1 appears, and no matter where the tag is placed on the automobile,the antenna is directed so the reading of the tag will take place withina maximum of 15 feet of the antenna 371. Assuming the antenna is aimedat the angle shown in FIG. 38, the tag 372 on the automobile in LANE 2will never be closer than about 25 feet from the antenna Accordingly,the signal sent back from that tag will be significantly smaller thanthe signal from any vehicle in LANE 1, as shown in the graph of FIG. 37for 25 feet and 15 feet, respectively. Therefore, the signal from theadjacent vehicle in LANE 2 will be well below the sensitivity level ofthe decoder/demodulator in apparatus of this invention used to extractthe vehicle identification in accordance with this invention. Vehiclesin LANE 1, however, will have almost a maximum signal strength whichwill be decoded very easily.

Active Transponder Second Embodiment

The modulated signal from a backscatter transponder such as described inthe first embodiment of the active transponder may also be detected bysensing edges of phase transitions of the received signal, indicative ofa change in modulation state. Thus, in the present second embodiment,the description of the interrogation signal transmitter, transponder,and symbol analyzer do not change. As in the first embodiment, thesignal from the stronger phase is preferably analyzed. These edges maybe detected by providing a differentiator or delay line 652, 653 with atime constant τ greater than the edge transition timing and less thanthe intersymbol timing and a change time constant for the interrogationsignal, and then detecting a phase transition edge. As noted above, themodulation scheme for typical backscatter RF-ID tags provides, for eachsymbol, a sequence of transitions. Thus, with each change in modulationstate, an accompanying edge is detected, which may be reconstructed intothe symbol sequence.

As shown in FIG. 44A, the receiving antenna 604 Rx introduces thereceived signals to a quadrature mixer 606, 607 for comparison in themixer 606, 607 with the interrogating RF signals from the source, or areference local oscillator 634, which, as stated above, may have anon-stationary frequency. The mixed quadrature signals are eachintroduced to an amplifier 608, 609 and a comparative signal strengthdetected using received signal strength indicator circuits 611, 613 andcomparator 614. In contrast to the first embodiment (and the embodimentdescribed below), these demodulated signals are not amplitude detectedfor sensing the bit pattern. Rather, the quadrature mixer 606, 607outputs are then each placed in a delay line 652, 653. Each quadraturemixer 606, 607 output is compared with a delayed representation thereof,with a delay of about 1-2 nS. The comparison may be of the analogsignal, using a mixer 654, 655, or as a digital comparison of a limitedsignal, in which case the delay line is preferably a shift register.When no phase transition is occurring, the output will be stable. Duringa transition, the delay will produce a disturbance in the output.Because the edge transitions are detected, the analysis is insensitiveto signal level. One of the outputs of the mixers 654, 655 is thenselected using circuit 659, and the edge transition stream converted toa bit stream with flip flop 658.

As shown in FIG. 44B, the transitions of the I and Q phase outputs ofthe mixers 606, 607 will be synchronized, but the respective signalamplitudes may vary, as determined by the RSSI 611, 613, as shown inFIG. 44C. The edge detector outputs 654, 655 therefore also vary inamplitude, as shown in FIG. 44D. The comparator 614 output is used toselect the stronger signal, through a selection circuit 659, which isthen passed through a flip flop 659 to produce a bit pulse as shown inFIG. 44E.

Dual Mode Transponder Interrogation System First Embodiment

The system according to the present invention is capable ofinterrogating and decoding an interrogation response from both passiveacoustic tags, active semiconductor tags, hybrids, and potentially othertypes. This is achieved by providing a non-stationary interrogationsignal within an interrogation band, as well as a receiver systemcapable of decoding both delay encoded characteristics of the receivedwaveform and sequentially modulated symbols of the received waveform.The receiver system therefore makes special accommodation to demodulatesequentially encoded symbols of the non-stationary carrier wave.

In its simplest form, a chirp (staircase frequency change) waveform RFsignal is emitted from the interrogation system. This signal istherefore directly compatible with the passive acoustic transpondersystem, and the essential circuitry remains relatively unchanged fromthat known in the art, for example as discussed above, using a homodynetype receiver. On the other hand, the chirp waveform poses certaindifficulties for interrogation of an active backscatter device. Inparticular, the phase change rate of the signal is relatively high, andtherefore the circuitry disclosed in U.S. Pat. No. 4,739,328 or4,888,591 will be ineffective for decoding sequentially modulatedsymbols of a non-stationary carrier, the phase of the carrier changingat too rapid a rate for the prior art circuitry to track and decode. Thecircuitry disclosed in U.S. Pat. No. 4,739,328, for example, whileemploying a four quadrant mixer 432, 434, provides for a signal polaritycomparison only, through limiters 446, 448, 450, XORs 466, 468, 470 andshift registers 454, 458, 462, with a shift register clock rate of fourtimes the frequency (one quarter the time period) of the secondfrequency (40 kHz), or about 160 kHz. This system does not respond toreceived signal amplitude, as this information is truncated in thelimiters 446, 448, 450. The third channel combiner 444 apparently sumsthe real and imaginary phases of the mixer output, thus producing acomposite signal. This known system however, does not compare thestrengths of the various phases, and the combined channel provides onlylimited information on the relative strengths of the I and Q signals.Therefore, noise in the signal, even if it instantaneously alters theamplitude of any signal component, remains unfiltered and may introduceerrors into the data analysis. At the limiter 446, 448, 450 stage, thedata is “digital”, and the circuitry does not address jitter or othertypes of artifacts which may appear in the limiter outputs.

The present invention demodulates both real and imaginary phases, andthen compares their amplitude to determine which has the greateramplitude, and presumably the greater signal to noise ratio. Theamplitude comparison occurs on a rectified and filtered portion of thesignal. This filtering allows various types of artifacts to be avoided.Further, the preferred embodiment oversamples the selected strongerphase to further allow noise detection and/or compensation.

The preferred system therefore mixes the received signal in a balancedmixer with quadrature phases representing the interrogation pulse, andthe mixed outputs low pass filtered to produce a pair of differencesignals. The signal strength of the difference signals are then assessedand the signals limited. The stronger signal at any given time is thenprocessed further, to decode the information contained therein. Wherethe signals are of comparable magnitude, either may be selected. Whereboth signals are below a threshold amplitude, the data may be ignored orflagged as potentially erroneous. The limited stronger signal is thensampled, for example, every 100 nS (10 MHz).

Due to the frequency change rate of the interrogation signal, as well asDoppler shift and round trip transmission delay of the interrogationpulse, the phase of the signal received from the antenna may changerapidly. With a chirp interrogation pulse, the difference between theemitted interrogation pulse and the reradiated signal will typically beless than about 3 kHz. This, however, is not the only possible type ofinterrogation signal. For example, a frequency hopping spread spectrumsignal may be used, which will have extremely high maximum frequencychange rates during hops. Since the hops are asynchronous with thesymbol transmission from an active tag, it is important to be able toquickly track the desired signal through a range of hops. In this case,a heterodyne receiver topology is preferred, with an intermediatefrequency (IF), for example 900 MHz, mixed with the return signal and ahopping frequency of between 5-25 MHz subsequently mixed with the outputof the IF mixer. During frequency hop transitions, it is likely that thedemodulator will generate artifacts, which may be identified and/orcorrected by appropriate processing according to the present invention.

By comparing the phase signal strength continuously and maintaining dataanalysis of the stronger phase, the present invention provides distinctadvantages.

In other embodiments, both I and Q phases may be analyzed, withconsideration of the comparative signal strengths, computer bit errorrates (BER), signal to noise ratio, or other metric of signal qualityemployed to weight any differences or disparities between the outputs.Further, the analysis is not limited to quadrature phases, and twophases having other than quadrature relationship may be analyzed, ormultiple phases, for example three phases each 60° apart may be,generated and analyzed. In the later case, the stronger one or twophases may be selected for analysis, or all or some of the phase signalsanalyzed with consideration of the comparative signal strengths,computer bit error rates (BER), signal to noise ratio, or other metricof signal quality employed to weight any differences or disparitybetween the outputs.

FIG. 49 shows a simplified schematic of a triple mode transponderinterrogation system: acoustic transponder, a first embodiment of thebackscatter transponder and a second embodiment of the backscattertransponder. A carrier oscillator 600 generates a stable carrier atabout 900 MHz. This carrier is mixed with the output of a non-stationarysignal generator 602 in a mixer 601. The non-stationary signal generatormay be, for example, a chirp generator, frequency hopping generator,direct sequence spread spectrum generator, digitally controlledoscillator, or other type of circuit. The mixed signal from the mixer601 is transmitted as a 905-925 MHz band signal from an antenna 604-Tx.The transmitted signal potentially interacts with different types oftransponders.

A first type includes an active modulator 623, which modulates theinterrogation signal received by the antenna 622 based on the output ofa state machine 624. A second type of transponder is a passivetransponder which subjects the signal received by the antenna 625 to aplurality of differing delays (τ) 627 a . . . 627 e, which are thensummed at node 626. In both types of transponder, the interrogationsignal is retransmitted as a reradiated signal to an antenna, forexample the antenna 604-Rx associated with the transmitting antenna604-Tx used to emit the interrogation signal.

The received signal is then mixed with a reference local oscillator 634output and an output of a quadrature phase generator 605, in mixers toproduce a set of quadrature “baseband” output signals from mixers 606and 607. The signals are buffered in buffers 608 and 609, and thenprocessed, for example with an NE624 (Signetics/Philips) integratedcircuit, which each include a limiter 610, 612 producing a limitedoutput 616, 617 and received signal strength indicator (RSSI) circuit611, 613. The received signal strength indicator signals 611, 613 arecompared by comparator 614, to determine which has a greater amplitude618. The received signal strength indicator signals 611, 613 are alsoseparately summed by network 621 and compared by a comparator 615 with avalue determined by another network 620 to produce a threshold signalstrength output 619.

The outputs of mixers 606 and 607 are also buffered with buffers 630,631. The signals are then processed by a signal processor 632 to producesets of frequency f₁, f₂, . . . , f_(N), and phase Φ₁, Φ₂, . . . , Φ_(N)outputs, which are then further processed in a processor 633.

Typically, a system would provide only one backscatter tag processingsystem, i.e. outputs 616, 617 or output 658; however, both are shown inFIG. 49 for comprehensiveness. The present example thus does not employthe circuitry specifically related to generating the flip flop 658output.

FIG. 45 shows a typical set of waveforms from a backscatter transpondertag.

FIGS. 46-48 detail the operation of the reader system for the activetransponder in more detail. The received signal from the transponder ismixed in a quadrature mixer to generate I and Q phase signals, asdiscussed above. As shown in FIG. 46, the structures of which arerepeated for each phase, the phase signal is amplified and filtered in asignal processing section 640. The processed signal is then input into aprocessor 641, which is, in this case, an NE624 integrated circuit. Thecircuit is configured to produce as an output a limited signalrepresenting the input as well as a received signal strength indicator(RSSI) signal, which is produced by rectifying and filtering (detecting)the phase signal, to produce a signal which corresponds to the amplitudeof the received signal.

FIG. 47 shows in more detail a circuit 642 for determining the strongerof the two phases, as well as a circuit for determining the compositesignal strength to determine whether it exceeds a threshold 643.

FIG. 48 shows schematically a logical circuit which analyzes the datastream, which includes the limited I and Q signals 650, 652, thecomparative magnitude signal 651 and the signal strength thresholdsignal 653. A 10.24 MHz clock 654 clocks a set of 7-bit counters 655,656, 657, 658 which act on I, I, Q, and Q, respectively. The high orderbit of each counter 655, 656, 658 is fed to a S input of an R/S latch659, 660, 662, respectively, with the high order bit of the counter 657fed to the S input of R/S latch 661. OR gates 663 and 667 receive thehigh order bit outputs of counters 655 and 656, and 657 and 658,respectively. OR gates 664 and 666 receives the Q outputs of R/S latches659 and 660, and 661 and 662, respectively. The penultimate order bit ofeach counter 655, 656, 657, 658 are together ORed in an OR gate 665. TheR input of R/S latch 659 and the R input of R/S latch 660 receives the Iinput; while the R input of R/S latches 661 and 662 receives the Qinput.

The output of quad input OR gate 665 serves as the clock to a D-typelatch 678, which receives the comparative magnitude signal 651 as the Dinput. The Q output of D-type latch 678 serves an a non-inverted inputto AND gates 668 and 679, and as an inverted input to AND gates 680 and681. The outputs of OR gates 663, 664, 666 and 667 serve asnon-inverting inputs to AND gates 668, 679, 680 and 681, respectively.The outputs of AND gates 668 and 681 are ORed in OR gate 669, and theoutputs of AND gates 679 and 680 are ORed in OR gate 682. The output ofOR gate 669 serves as a clock input to shift register 671, which, inturn, provides a parallel output to a P-term matrix 672. The output ofOR gate 682 passes through a 1-bit counter 670, whose output serves as adata input to the shift register 671.

The data output is then generated by AND gate 683, OR gate 684, D-typelatches 673 and 677, XOR gate 685, and counter 676, based on the P-termmatrix 672 output, OR gate 669 output, and the signal strength thresholdsignal 653, the later of which blocks output when the signal strength istoo low through a clear input of D-type latch 673, as shown in FIG. 48.This circuit thus recovers both Data 674 and Data clock 675, for use bya processor.

The P-term matrix is implemented in a PLD, with the following formulas:Zero's=/((Q ₀ · Q ₁ ·Q ₂ · Q ₃ ·Q ₄ · Q ₅)−( Q ₀ ·Q ₁ · Q ₂ ·Q ₃ · Q ₄·Q ₅))One's=/((Q ₀ · Q ₁ · Q ₂ ·Q ₃ ·Q ₄ · Q ₅−( Q 0·Q ₁ ·Q ₂ · Q ₃ · Q ₄ ·Q₅))

The detection strategy operates using the P-term matrix to detect “half”bits (1's, 0's), by detecting a 1 0 pattern, indicative of a zero, a 0 1pattern, indicative of a one, or a synchronization pattern 1 1 1 0 or 00 0 1. The clock is recovered from the data edges in a digital clockrecovery system.

Appendix A provides sample assembler and C language code for a routinefor collecting the data from the digital processor for analyzing the tagsimilar to that disclosed in U.S. Pat. No. 4,739,328, and available fromAmtech Corp., Santa Fe, N. Mex.

Dual Mode Transponder Interrogation System Second Embodiment

The first embodiment of the dual mode transponder interrogation systememploys an received signal strength indicator (RSSI)-responsiveselection of signal component for detection (e.g., analysis of signalamplitude information) of backscatter transponder information. Incontrast, the second embodiment of the dual mode transponderinterrogation system employs the modulation state detector of the secondembodiment of the active transponder system, a phase change sensitive(“edge”) detector. Otherwise, the systems are similar. The presentexample thus employs the circuitry to generate the flip flop output 658,which in turn corresponds to the stronger phase signal of the firstembodiment, and may be analyzed to determine the modulation informationaccordingly.

The preferred system therefore mixes the received signal in a balancedmixer with quadrature phases representing the interrogation pulse, andthe mixed outputs low pass filtered to produce a pair of differencesignals. The signal strength of the difference signals are thenassessed. The stronger signal at any given time then controls theprocessing of a corresponding data stream which is the output of thequadrature mixers passed through an edge detector, to produce transitioninformation. This transition information then be reconstructed to a bitstream through a flip flop, and processed further, to decode theinformation contained therein. Where the signals are of comparablemagnitude, either edge detection stream may be selected. Where bothsignals are below a threshold amplitude, the data may be ignored orflagged as potentially erroneous. The limited stronger signal is thensampled, for example, every 100 nS (10 MHz).

As shown in FIG. 49, the signal received by the receiver is mixed with areference local oscillator 634 output and an output of a quadraturephase generator 605, in mixers to produce a set of quadrature “baseband”output signals from mixers 606 and 607. The signals are buffered inbuffers 608 and 609, and then processed, for example with an E624(Signetics/Philips) integrated circuit, which each include a limiter610, 612 producing a limited output 616, 617 and received signalstrength indicator (RSSI) circuit 611, 613. The received signal strengthindicator signals 611, 613 are compared by comparator 614, to determinewhich has a greater amplitude 618. The received signal strengthindicator signals 611, 613 are also separately summed by network 621 andcompared by a comparator 615 with a value determined by another network620 to produce a threshold signal strength output 619.

The outputs of mixers 606 and 607 are also buffered with buffers 630,631. The signals are then processed by a signal processor 632 to producesets of frequency f₁, f₂, . . . , f_(N) and phase Φ₁, Φ₂, . . . Φ_(N)outputs, which are then further processed in a processor 633.

The outputs of mixers 606 and 607 are further buffered with buffers 650,651. The buffered signals each then pass through an edge detector,having a 2 nS analog delay line 652, 653 and a double balanced mixer654, 655 comparing the delayed and undelayed representations of eachphase signal. The double balanced mixer 654, 655 outputs are thenanalyzed with circuit 656, 657 for an impulse, indicative of a phasetransition. The impulse detection circuit 656, 657 output correspondingto the stronger signal phase is then selected based on the comparator614 output. The selected impulse detection circuit output is thenconverted into a bitstream with flip flop 658, and further analyzed bythe signal analyzer, as described above and shown in FIG. 48.

There has thus been shown and described a novel RF-ID tag interrogationsystem, which fulfills all the objects and advantages sought therefor.Many changes, modifications, variations and other uses and applicationsof the subject invention will, however, become apparent to those skilledin the art after considering this specification and the accompanyingdrawings which disclose preferred embodiments thereof. All such changes,modifications, variations and other uses and applications which do notdepart from the spirit and scope of the invention are deemed to becovered by the invention which is limited only by the claims whichfollow.

1. An RFID reader, comprising: a transmitter configured to transmit RFsignals to a plurality of RFID tags configured to modulate the RFsignals according to first and second modulation schemes, said pluralityof RFID tags including a first plurality of backscatter modulation tagsconfigured to backscatter modulate RF signals from said transmitteraccording to said first modulation scheme and a second plurality ofbackscatter modulation tags configured to backscatter modulate RFsignals from said transmitter according to said second modulationscheme; a receiver including a demodulator configured to receive anddemodulate RF signals backscattered from said plurality of RFID tagsthat are modulated according to the first and second modulation schemes;and a decoder configured to decode received signals that are encodedaccording to different encoding schemes.
 2. The RFID reader of claim 1wherein said transmitter includes a spread spectrum generator and saidtransmitter is configured to generate and transmit spread spectrum RFinterrogation signals to the plurality of RFID tags.
 3. An RFID readerof claim 1 wherein said demodulator is configured to demodulatephase-shift keying modulated signals.
 4. The RFID reader of claim 3wherein said demodulator is configured to demodulate amplitude-shiftkeying modulated signals.
 5. An RFID reader, comprising: a transmitterconfigured to transmit RF signals to a plurality of RFID tags; areceiver including a demodulator configured to receive and demodulate RFsignals reflected from said plurality of tags and modulated according tofirst and second modulation schemes; and a decoder configured to decodesignals from said plurality of tags that are encoded according todifferent encoding schemes.
 6. The RFID reader of claim 5 wherein theplurality of RFID tags includes a first plurality of backscattermodulation tags from which RF signals modulated according to the firstmodulation scheme are reflected, and said demodulator is configured todemodulate RF signals reflected from the first plurality of backscattermodulation tags.
 7. The RFID reader of claim 6 wherein the plurality ofRFID tags further includes a second plurality of backscatter modulationtags from which RF signals modulated according to the second modulationscheme are reflected, and said demodulator is configured to demodulateRF signals reflected from the second plurality of backscatter modulationtags.
 8. The RFID reader of claim 5 wherein said transmitter includes aspread spectrum generator and said transmitter is configured to generateand transmit spread spectrum RF interrogation signals to the pluralityof RFID tags.
 9. An RFID reader of claim 5 wherein said demodulator isconfigured to demodulate phase-shift keying modulated signals.
 10. TheRFID reader of claim 9 wherein said demodulator is configured todemodulate amplitude-shift keying modulated signals.
 11. An RFID reader,comprising: a transmitter configured to transmit RF signals to a firstbackscatter modulation tag configured to backscatter modulate RF signalsaccording to a first modulation scheme and to a second backscattermodulation tag configured to backscatter modulate RF signals accordingto a second backscatter modulation scheme; and a receiver including ademodulator configured to demodulate RF signals backscattered from thefirst backscatter modulation tag and demodulate RF signals backscatteredfrom the second backscatter modulation tag.
 12. The RFID reader of claim11, further comprising a decoder configured to decode received signalsthat are encoded according to different encoding schemes.
 13. The RFIDreader of claim 11 wherein said transmitter includes a spread spectrumgenerator configured to generate and transmit spread spectrum RFinterrogation signals to the first and second tags.
 14. The RFID readerof claim 11 wherein the first tag is configured to backscatter modulateusing phase-shift keying (PSK) and said demodulator is configured todemodulate PSK signals.
 15. The RFID reader of claim 14 wherein thesecond tag is configured to backscatter modulate using amplitude-shiftkeying (ASK) and said demodulator is configured to demodulate ASKsignals.